5 ! l

FOM-lnstituut voor Plasmafvsica 'Riinhuizen'

Associatie EURATOM-FOM. Edisonbaan 14. 3439MN Nieuweaein. Nederland

Abstract We present a relatively low cost 50 MW modulator

capable of delivering a 240 kV pulse with a pulse top flatness within 0.05 Yo. The duration of the flat top is 20 ps. The pulse is generated by a pulse forming network (PFN), a thyristor switch and a step-up transformer. The PFN, designed for 40 kV charging voltage, consists of four parallel lines, each containing only 19 capacitors and 18 inductors. An innovative line switch is used, namely a stack of 32 thyristors in series. The step-up transformer has an iron core and its voltage ratio is 1 to 12.

Introduction

The modulator is used to provide the cathode voltage for a klystron, which supplies 20 MW of rf power at 3 GHz for a rf linac based Free Electron Laser. [l] The extremely good voltage stability is needed to minimize the energy variation of the electron beam, which otherwise would strongly deteriorate the laser performance. We have chosen a modulator composed of a switched-mode power supply, a PFN, a thyristor switch and a pulse transformer. A simplified circuit diagram of the modulator is given in Fig. 1. The maximum charging voltage for the PFN is 40 kV which permits a modulator layout in air for easy maintenance. A thyristor switch is preferred over a thyratron because it has a more stable voltage drop. The pulse transformer and the base of the klystron with their associated components are immersed in oil.

m Main performance requirements

Output voltage -240 kV Output power 50 MW Maximum pulse duration 25 PS Flat top duration 20 PS Stability of flat top 0.08 yo Repetition frequency 10 Hz

Numerical simulations were performed in order to investigate the minimum ripple obtainable. Also the influence of capacitor tolerances was investigated. Instead of using PSpice, all simulations were done with a home made programme written in Pascal for a Macintosh Plus computer.

PFN (end) 4x 1 9 c t i o n s

The reason for this is that a routine for optimization could be added which modifies inductor values to obtain the best pulse-top flatness. A second reason is that PSpice needs a fixed time step in order to obtain the required precision which makes it much slower than a dedicated programme. There was no difference between the outputs of both codes. In all PFN simulations, the PFN is connected to a complex load impedance representing the expected values of the transformer and the klystron at full power. Non-linearities of klystron and pulse-transformer are ignored and the switch is supposed to be ideal. A single line is used for the PFN, representing the four lines in parallel. The nominal values for the simulations of the PFN components and the load are shown in Fig. 2.

PFN LOAD

90 nF

Fig. 2. The nominal values of the PFN and the load

The first simulation shows the output for an optimized PFN with nominal capacitor values. The pulse top is shown in Fig. 3a. The voltage at C10 in Fig. 3b shows the discharge step from 40 to 20 kV, followed by an oscillation.

In the second simulation, capacitor C10 was given a 10 % higher value. Fig. 4a shows that the distortion of the pulse consists of two parts. Firstly there is a reflection due to impedance mismatch and secondly an oscillation follows the reflection. This is the same oscillation which shows up in Fig. 3b. If the propagation time in every section of the PFN is constant, these oscillations cancel out each other at the load very effectively as can be seen in Fig. 3a. After the inductors adjacent to C10 were given a 5 Yo higher value (in order to restore the characteristic impedance in the PFN), the reflection has disappeared but the oscillation has become worse, see Fig. 4b. This is due to the fact that impedance scales as d(L/C) whereas propagation time scales as d(LC). It is clear that impedance mismatch induces reflections and propagation-time mismatch induces oscillations. Because there is always a tolerance spread in real capacitors, the capacitors for our PFN were sorted on value and mounted in an up-going order from the front-end of the PFN to avoid discontinuities. As the real PFN has four lines in parallel, the capacitor values of corresponding sections in each line were added and the sum is used in the third simulation.

(front end)

Fig. 1. Simplified circuit diagram of the modulator

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CH3180-7/92/0000-0123$1 .OO 0 1992 IEEE

0.01 I I I I I I I

t . 0.00

- E

-0.01

25 0 0 1 5 10 15 20 25 30

time (PSI

Fig. 3. Low voltage simulation for an optimized PFN with nominal capacitor values: (a) top of the pulse, (b) the voltage on C10

21 I I 1 I I 1

% 0.4 r ' 0.2

0.0

-0.2

-0 .4 0 5 10 15 20 25 30

time (ps)

Fig. 4. (a) Output distorted by a deviation of value of C10. (b) the PFN corrected for constant impedance.

-0 * 02 I I I I I I I I I 0 5 10 15 20 25 30

time (ps)

Fig. 5. Top of the output pulse of the optimized PFN with capacitor values of the real capacitors.

In the third simulation Fig. 5 was obtained after the PFN had been optimized. The ripple is about 0.02 Yo, which is only two times as high as in the first simulation, see Fig. 3a.

The thvristor switch

We have chosen the innovative approach of using a stack of 32 reverse-conducting thyristors (ABB, type CSR331-20) as switch. The total hold-off voltage of the 32 thryristors in series is 57 kV. Each thyristor has its own gate transformer circuit, voltage divider resistor with LED, snubber and a punch-through voltage protection circuit with break-over diodes. The gate pulse generator has a number of built-in control circuits to ensure that the gate current pulses are sufficiently high and long, since too low or too short pulses would cause thyristor damage. Special care was taken to avoid corona discharges because they can lead to unintentional thyristor firing. During a test of half an hour at 60 kV DC, no corona discharges were found. A saturable inductor has been placed between the thyristors and the pulse transformer in order to protect the thyristors against the large dl/dt-values that otherwise would develop in case of a breakdown. This inductor blocks the 40 kV voltage during 0.6 ps after firing the thyristors. The magnetizing current is 100 A. After 0.6 ps, the thyristor conductivity has become sufficiently large to withstand a breakdown. [2] The residual inductance after saturation of the ferrite core is 6 pH.

The pulse transformer

The main characteristics of the transformer, custom-built by Pearson, are given in Table 2.

I&l!s Charactenstics of the transformer . .

Voltage ratio 1 to12

Maximum pulse duration 20 PS Maximum droop 2 YO Pre-magnetizing current 11 A

Output voltage 240 kV

The core of the transformer consists of two joined "U" parts. On each leg a primary and a secondary winding is mounted. The two primaries are connected in parallel. Ttie heater power for the klystron is fed through the secondaries at a voltage level of 220 V AC and converted to a smoothed 28 V DC. For the HV pulse the two secondaries are also connected in parallel but they carry the heater current for the klystron in series. Although in this way, the heater current circuit is very well decoupled from the pulse current circuit, it is a discomfort to feed AC through the secondary, because the heater current produces magnetic fields in both secondary coils. The fields are counteracting for the closed magnetic loop of the core, but leave the core at the rounding of the "U" parts. This external stray field perturbes the cathode current of the klystron.

Another difficulty is that the secondary windings form a parasitic "delay line" because they have distributed stray capacitance and stray inductance. At the start of the pulse, .a standing wave is generated because the "delay line" IS shorted at both ends. Its characteristic impedance is about 5 times higher than the klystron impedance. Hence, the maximum standing wave voltage is half way the secondary windings and currents flow in opposite direction. This implies that this current flows through the klystron cathode but cannot be seen at the primary side of the transformer. The wave must be damped at the HV side. One effective HV damping circuit has been connected to both secondary windings, see Fig. 6. Two taps were installed halfway these windings. The circuit was adjusted during operation.

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Fxperimental results

Initially the PFN was tuned line by line, charging it to 25 V and switching it on a dummy load by means of a low- voltage power FET. The dummy load was composed of low voltage electronic components with four times the impedance as given in Fig. 2. After connecting them in parallel, a dummy load with values as given in Fig. 2 was used for a low voltage measurement. The output pulse and the top are shown in Fig. 7.The flatness is in the order of 0.02 Yo. The oscillation of about 1 MHz was caused by the inductance of the busbar connecting the four lines at the front-end of the PFN and the capacitors distributed along the busbar. The oscillation was eliminated by placing the capacitors close together and connecting the out-going HV cable at these capacitors.

The next step was to start HV operation. The thyristor switch, the transformer and the klystron were connected. As a HV divider with the required precision was not available, all high voltage output signals were derived from the phase-shift at 3 GHz between input and output of the klystron. This method requires a 3 GHz input signal with a low phase-jitter. The conversion factor is 0.15 % per degree and depends on the mechanical layout of the klystron and the velocity of its electron beam. At the front-end of the PFN an extra capacitor of 60 nF had to be added to make up for the eddy-current losses of the transformer. The top of the pulse revealed the expected droop of the transformer and an oscillation caused by the "delay line" effect which was mentioned earlier. Fig. 8 shows the oscillations and the droop of the transformer on top of the pulse. After the damping circuit of Fig. 6. was installed, the PFN and the damping circuit were tuned for optimum pulse flatness. The fundamental frequency of the standing wave has almost disappeared, see Fig. 9. Only a higher frequency is still present but this frequency is beyond the cut-off frequency of the accelerator. The stability is in the order of 0.05 Yo at the start of the pulse and much better during the last 15 ps.

Damping circuit hHVh 1.

'I1 200kV - I + Fig. 6. The damping circuit for the pulse transformer.

,

Fig. 7. The output pulse of the PFN on a dummy load at 25 V charging voltage: (A) the full pulse, (B) the expanded top.

Fig. 8. The top of the output pulse of the modulator for the initially flat top as in Fig. 7, distorted by the transformer.

Fig. 9. The final output pulse of the modulator which has a ripple of about 0.05 Yo.

Conclusion.

It is possible to build a modulator with a precision of 0.05 % with only 19 sections in the PFN which considerably reduces the costs. The total cost of the described modulator (including salaries) amounts to about k$ 500.

Acknowledaements The advice of E. Heine and F.B. Kroes (NIKHEF-K,

Amsterdam), C.R. Dunbar and J.B. Lyall (SERC Daresbury Laboratory, Daresbury), and M.G. Kelliher (Kelvin Laboratory, Glasgow) in designing our modulator has been indispensable and is gratefully acknowledged. We are indebted to A.M. van lngen and A.B. Sterk of our Institute for valuable discussions and assistance. This work has been performed as part of the research programme of the association agreement between the Stichting voor Fundamenteel Onderzoek der Materie (FOM) and EURATOM, with financial support from the Nederlandse Organisatie voor Wetenschappelijk Onderzoek (NWO) and EURATOM.

1.

2.

References,

Update on FELIX, P. W. van Amersfoort et al. Nucl. Instr. and Meth. A304 (1991) 163.

Fast Switching Thyristors Replace Thyratrons in High- Current Pulse Applications. Publication No. CH-EC 1661 87 E. ASEA BROWN BOVERI. A. Schweizer, J.L. Steiner, J. Vitins.

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