Post on 02-Feb-2020
TMTT-2015-01-0083
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Abstract—This paper proposes a general synthesis method for
inline mixed coupled quasi-elliptic bandpass filters based on λ/4
resonators. Different from the conventional modeling with
lumped and mixed inductive and capacitive coupling, the
proposed mixed-coupling model here includes not only the
coupling magnitudes, but also their paths’ phases inside the
mixed-coupling. According to the presented formulation, there
exists one lower side transmission zero for the capacitive-
dominant mixed coupling. However, there are two transmission
zeros in the upper stopband for the inductive-dominant case, and
these two transmission zeros can be analytically synthesized with
different coupling path ratios. The proposed synthesis method is
then applied to design two 2nd-order bandpass filters based on
inductive-dominant coupled λ/4 resonators, resulting in two
upper stopband transmission zeros. Furthermore, this synthesis
method is extended to design two 4th-order quasi-elliptic
bandpass filters with both inline capacitive- and inductive-
dominant mixed couplings. Finally, the designed filters are
fabricated and measured to provide successful verification on the
proposed mixed coupling model approach.
Index Terms—Inline coupled filter, mixed coupling, quarter-
wavelength resonator, quasi-elliptic bandpass filter and finite
transmission zero.
I. INTRODUCTION
INITE transmission zeros are critical for filter synthesis to
enhance the rejection ratios. One of prevalent methods for
creating finite transmission zeros is the basic cross-coupling
mechanism [1]. These cross couplings are equal in magnitude
and out-of-phase with the mainline one at the transmission zero
frequencies. Cross-coupling paths can be introduced either by
resonant nodes [2]–[4], or non-resonant nodes [5], [6]. This
mechanism is only valid with assumption of narrow-band
non-dispersive mainline and cross couplings, and it is able to
generate maximum N-2 finite transmission zeros without
considering I/O ports (N is the filter order). Another popular
mechanism to create finite transmission zeros is based on the
mixed inter-resonator couplings, where capacitive- or
inductive-dominant mixed-coupling is dispersive. In the filter
core-passband synthesis, it provides required capacitive- or
inductive-dominant couplings. Meanwhile, these capacitive
Manuscript received November 7, 2014; revised January 23, 2015, May 8,
2015 and August 3, 2015; accepted on August 5, 2015.
S. Zhang and R. Weerasekera are with Institute of Microelectronics, Singapore. S. Zhang is also with Skyworks Solutions, Singapore. (e-mail:
songbai.zhang@outlook.com)
L. Zhu is with the Department of Electrical and Computer Engineering, Faculty of Science and Technology, University of Macau, Macau SAR, China.
and inductive components annihilate with each other at certain
frequencies, and resulting in finite transmission zeros.
Mixed-coupling between open-loop λ/2 resonators was
discussed in [7], and its inductive and capacitive components
are enlaced with each other. An earlier work [8] employed the
stepped impedance resonators (SIRs) to separately control
inductive and capacitive components in mixed inter-resonator
couplings. Separated inductive coupling using iris and
capacitive coupling using strips were also introduced between
two cavity resonators [9] and the substrate integrated
waveguide (SIW) resonators [10]–[12] for implementation of
compact inline quasi-elliptic bandpass filters. For the other
planar implementations, quasi-elliptic bandpass filters based
on λ/4 resonators and separated J and K mixed-couplings were
illustrated in [13]–[18]. However, most of pioneered works had
in general modeled the mixed-coupling as in Fig.1 (a) with a
lumped inductor, Lm, and a capacitor, Cm, intuitively
representing for inductive and capacitive couplings,
respectively. Explicit formulation presented in [7], [11] had
explained that existence of one lower-side transmission zero
for the capacitive-dominant coupling, and one upper-side
transmission zero for the inductive-dominant coupling.
Synthesis of Inline Mixed Coupled Quasi-Elliptic
Bandpass Filters Based on λ/4 Resonators
Songbai Zhang, Member, IEEE, Lei Zhu, Fellow, IEEE, Roshan Weerasekera, Senior Member, IEEE
F
Cm
Lm
C C
L L
(a)
J
K
θj
θk
θ1,
θ2,
θu
θl
θj
θk
θu
θl
Port 1 Port 2
Zr
Zr
θ1, Zr
θ2, Zr
(b)
Fig. 1. (a) Conventional lumped model for mixed inter-resonator coupling, (b)
proposed mixed coupling model for λ/4 resonators with coupling path phases.
TMTT-2015-01-0083
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To the best of our knowledge, for the first time in this paper,
mixed-coupling is modeled not only with magnitudes of
constituted capacitive and inductive couplings, but also their
respective different coupling path phases as illustrated in Fig
1(b). Compared with the conventional model in Fig. 1(a), there
are two coupling paths between ports 1 and 2 in Fig. 1(b). The
upper path composes of two 𝜃1 sections, a J inverter and its
two associated electric lengths 𝜃𝐽. On the lower path, there are
two 𝜃2 sections, a K inverter and its two associated electric
lengths 𝜃𝐾 . Overall Fig. 1(b) portrays that two λ/4 resonators
are mixed coupled together with separated J and K inverters.
Depending on different positions of ports 1 and 2, there exists
different phase difference between these two coupling paths.
By introducing this coupling path with phase difference, a
single lower side finite transmission zero appears for the
capacitive-dominant inter-resonator mixed coupling, which is
identical with the conventional model in Fig. 1(a). Different
from lumped modeling in Fig. 1(a), two upper-side
transmission zeros emerge for the inductive-dominant
mixed-coupling case in Fig 1(b), and the pertained pair of
transmission zeros can be analytically derived and controlled
with different choices of coupling path phases.
The remainder of this paper is organized as follows. In
Section II, in-band mixed-coupling for filter synthesis is
discussed first. Transmission zeros due to the mixed-coupling
are formulated, and then mathematically proved the existence
of one lower-side and two upper-side transmission zeros for the
capacitive-/inductive-dominant mixed couplings, respectively.
In Section III, two 2nd
-order planar bandpass filters with
inductive-dominant mixed-coupled λ/4 resonators are designed
to experimentally verify two upper-side transmission zeros.
This synthesis methodology is further extended to design two
4th
-order inline quasi-elliptic filters in Section IV. Lastly,
Section V concludes this paper.
II. MIXED COUPLING FORMULATION
A. In-Band Mixed Inter-Resonator Coupling
As shown in Fig. 1(b), two λ/4 resonators are simultaneously
coupled with separated J and K inverters. These two inverters,
represent electrical and magnetic couplings, are out-of-phase
for the passband synthesis [19]. The overall inter-resonator
coupling coefficient 𝑀𝑖𝑗 is calculated in following (1).
b
J
x
KFBW
ijm
ijM (1)
where 𝑚𝑖𝑗 is the normalized entry in coupling matrix, FBW is
the fractional bandwidth. K and J are the inductive and
capacitive inverters, their signs are rather relative. Inductive
inverter K is defined positive hereafter, and the capacitive J is
thus chosen as negative.𝑥 = 𝜋𝑍𝑟 4⁄ and 𝑏 = 𝜋𝑌𝑟 4⁄ are λ/4
resonator’s reactance and susceptance slopes, respectively [3].
𝑍𝑟 and 𝑌𝑟 are the characteristic impedance and admittance of
λ/4 resonators in Fig. 1(b). (1) can be further simplified as (2),
) ˆˆ4)(
4JK(
Y
J
Z
KM
rr
ij
(2)
Here, the normalized J and K inverters are defined as
rZKK /ˆ andrYJJ /ˆ . J and K inverters can then be
synthesized based on our previous work in [20] for the physical
realization.
B. Mixed Coupling and Finite Transmission Zeros
In Fig. 1(b), the upper path is coupled with the capacitive J
inverter, and lower path is coupled with the inductive K
inverter. ABCD matrix for the upper capacitive coupling path is
calculated in (3) at the bottom of this page. Admittance matrix
element 𝑌21,𝑢 for the upper network is thus calculated in (4),
uur
uZJ
jJY
2222,21cossin
(4)
Meanwhile, ABCD matrix for the lower half inductive coupling
path is calculated in (5) at the bottom of this page.𝑌21,𝑙 in
admittance matrix is derived in (6).
llr
lKZ
jKY
2222,21cossin
(6)
The upper capacitive and lower inductive coupling paths are
connected in parallel, so the overall normalized 𝑌21 with
respective to resonator’s admittance 𝑌𝑟 between ports 1 and 2
in Fig. 1(b) is calculated as follows.
r
lu
r Y
YY
Y
YY
,21,212121
ˆ
uurJZ
J
rY
ujJ
J
urjY
uJ
rjZ
J
uj
uuJ
rZ
J
rY
uurjY
urjZ
ujJ
Jj
uurjY
urjZ
u
UABCD
cossin)(2cos
2sin2
2sin22cos
cossin)(
cossin
sincos
0
/0
cossin
sincos
(3)
llrr
lrl
llr
llrr
llr
lrl
llr
lrl
L
KYK
ZKjY
K
j
jKK
jZKY
K
Z
jY
jZ
Kj
jK
jY
jZ
ABCD
cossin)(sincos
cossin
cossin)(
cossin
sincos
0/
0
cossin
sincos
222
222
(5)
TMTT-2015-01-0083
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llK
Kj
uuJ
Jj
2sin2cos2ˆ
ˆ
2cos2sin2ˆ
ˆ
(7)
To create transmission zeros, magnitude of 21Y should be
zero, thereby its numerator in (8) is equal to zero.
)ˆ(numerator ˆ2121 |Y|Y ,n
uK
uKJ
lJ
lKJ 2cosˆ2sinˆ2ˆ2sinˆ2cos2ˆˆ (8)
Before analyzing (8) in general, a special case is worthy of our
attention. Assuming that (a) both J and K inverters are
non-dispersive, and (b) the electrical lengths of upper and
lower coupling paths in Fig. 1(b) are equal to each other, e.g.
u = l = 0 . Thus, transmission zeros are constrained as
02tan
ˆ
ˆ
J
K (9)
Fig. 2 illustrates transmission zeros configuration. For a λ/4
resonator filter with central frequency 0f locates at 4/0 ,
its 1st spurious response appears around 03 f where 4/30 .
Thus, only transmission zeros below 03 f are discussed
hereafter. According to (9), if the mixed coupling is
capacitive-dominant, i.e., KJ ˆˆ , the mixed coupling in (2)
becomes negative. As a result, a single transmission zero
appears at lower side of passband in 1ˆ/ˆ JK region in Fig. 2.
However, if the mixed coupling is inductive-dominant, in-band
coupling in (2) is positive. Accordingly, two transmission
zeros appear in 1ˆ/ˆ JK region in Fig. 2, and they are
symmetrical with respect to 2/0 . Now, analysis here
proves existence of two upper-side transmission zeros for the
inductive-dominant mixed coupling. However, both practical J
and K inverters are dispersive, and positions of two ports in
Fig. 1(b) are also not restricted in middle of λ/4 resonators.
Hence, exact transmission zeros positions are subjected to the
general condition in (8), which will be discussed further in the
following section.
III. MIXED-COUPLED 2ND
-ORDER FILTERS
Based on the previous discussion, two inductive-dominant
2nd
-order Chebyshev bandpass filters are exemplified with
different port locations. Both filters are designed with 20 dB
in-band return loss with fractional bandwidths of 3.7%, and
central frequency of 2.5 GHz. The dielectric substrate used in
this work is Roger’s RT/Duriod 6010 with εr=10.8, tanδ =
0.002 and thickness = 1.27 mm. The strip width of λ/4
resonators is uniformly chosen as 0.4 mm, and its characteristic
impedance corresponds to 𝑍𝑟 =73.7 Ω. The coupling
coefficient M12 based on 2nd
-order Chebyshev lowpass
prototype is presented in (10),
061.05445.06648.0
%7.3
21
12
gg
FBWM
(10)
Hence, the required mixed coupling is derived according to (2),
048.04
061.04
ˆˆ12
MJK (11)
The J inverter in Fig. 1(b) is realized with the anti-parallel
coupled line in Fig. 3(a). A quasi-lumped metallic via in Fig.
3(b) functions as the K inverter in Fig. 1(b). Their physical
dimensions are explicitly illustrated in Fig. 3(a) and 3(b),
respectively. Their broadband normalized values: J and K , and
their associated electrical lengths: 𝜃𝐾 and 𝜃𝐽 , are fullwave
extracted and shown in Fig. 3(c) [20]. From Fig. 3(c), K
=0.108, J =0.06, 𝜃𝐾 =7.660, and 𝜃𝐽 =19.05
0 at 2.5 GHz. This
inductive-dominant mixed coupling calculated in (11) satisfies
the theoretical inter-resonator coupling requirement M12=0.061
in (10). Furthermore, the positions of two upper-side
Fig. 2. Transmission zeros configuration emerged in presented filter based on
mixed and equal-phase coupled λ/4 resonators.
L=2.0 mm
S=0.7 mm
W=0.4 mm
de-embed de-embed
Zr=73.7 Ω Zr=73.7 Ω
de-embed de-embed
Wstub=0.6 mm
Lstub=0.9 mmLvia=0.6 mm
D=0.4 mm
Zr=73.7 Ω Zr=73.7 Ω
(a) (b)
(c)
Fig. 3 Schematics of (a) an anti-parallel coupled J inverter, and (b) a
quasi-lumped K inverter. (c) Fullwave extracted normalized J and K inverter
values and associated electrical lengths.
TMTT-2015-01-0083
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transmission zeros are determined based on (8) and Fig. 3(c).
Fig. 4 (a) and (b) graphically illustrate two upper-stopband
transmission zeros with different k, which is defined as
electrical length ratio of upper to lower paths in Fig. 1(b).
l
uk
(12)
Fig. 4(a) depicts the graphs of normalized admittance for
determining the transmission zeros with k ranged from 0.42 to
0.56. The pair of transmission zeros emerge at k=0.43, and split
afterwards. As k increases from 0.43 to 0.56, the first
transmission zero fZ1 shifts downwards, and 2nd
transmission
zero fZ2 shifts more aggressively to the higher frequency. As k
increases above 0.7 as plotted in Fig. 4(b), transmission zero fZ1
continues to slightly shift downwards. However, transmission
zero fZ2 shifts backward to lower frequency. In this context,
these two upper transmission zeros can be derived and
controlled with different port positions in Fig. 1(b).This result
also addresses the two upper transmission zeros in [15], [18].
To realize the external coupling structure between I/O ports
and resonators, the tapped line structure proposed in [21] is
applied as used in [13]. However, the I/O tapped line positions
(ports 1 and 2’s positions in Fig. 1(b)) are primarily determined
by required external quality factor 𝑄𝑒, thus, it results in limited
freedom for controlling length ratio k, and associated two
upper-side transmission zeros. To circumvent this problem, a
tri-port coupled-line in Fig. 5(a) is used. Its equivalent circuit
was developed based on [22], and illustrated in Fig. 5(b).
Electrical length in Fig. 1(b) is determined by coupled line
length 𝜃 , and external quality factor 𝑄𝑒 depends on the
coupling gap and transformer ratio n in Fig. 5(b). Hence, length
ratio k and external quality factor 𝑄𝑒 can be de-coupled and
separately designed.
These two 2nd
-order bandpass filters, noted as Filter A and
Filter B, are shown in Fig. 6(a) and 6(b), respectively. Both
filters have identical in-band frequency responses; therefore,
(a)
(b)
Fig. 4. Admittance graphs under different length ratios for determining
transmission zeros (a) 56.042.0 k , and (b) 9.15.0 k .
Port 1 Port 2
Port 3
Port 1 Port 3
Port 2
θ θ
1:n
(a) (b)
Fig. 5. (a) Tri-port coupled line section, and (b) equivalent circuit [22] .
(a) (b)
(c)
(d)
Fig. 6. Physical filter layout with different coupling length ratios, (a) k=0.5,
and (b) k=0.45. Fullwave simulated and measured frequency responses for (c)
Filter A and (d) Filter B.
TMTT-2015-01-0083
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they have the same inductive-dominant coupling M12. The
constituted J and K inverters are illustrated in Fig. 3(a) and (b).
However, Filter A and Filter B have different length ratios
contributed by the tri-port feeding structure in Fig. 5(a). In our
design, the length ratios are chosen as 0.5 and 0.45 for the
Filter A and Filter B, respectively. According to the Fig. 4(a),
the theoretical transmission zeros appear at 3.5 and 6.8 GHz for
Filter A, 3.8 and 5.3 GHz for Filter B. Apart from the
associated electrical lengths 𝜃𝐽 and 𝜃𝑘 in relevance to J and K
inverters, compensated lengths 𝜃1=110 or 1.4 mm for upper
capacitive coupling path and 𝜃2=52.340 or 6.75 mm for the
lower inductive coupling path in Fig. 1(b) are introduced to
realize effective λ/4 resonators for Filter A with length ratio k
=0.5. Similarly, 𝜃1=8.90 or 1.15 mm and 𝜃2=54.4
0 or 7.0 mm
are inserted for upper and lower coupling paths in Filter B,
whose length ratio k=0.45, for the same purpose.
Fig. 6(a) and 6(b) show the optimized filter layout with noted
dimensions. Insets in Fig. 6(c) and 6(d) illustrate two identical
simulated in-band frequency responses for Filter A and Filter
B, and they are consistent with measured results. Measured
in-band insertion and return losses are 1.05 dB and 21.4 dB for
Filter A, and 0.90 dB and 17.6 dB for Filter B. The measured
transmission zeros are found at 3.47/6.77 GHz and 3.64/5.12
GHz for Filter A and Filter B, respectively. These measured
and simulated two transmission zeros locations for both filters
with different feeding length ratios have justified our proposed
idea in the design.
IV. MIXED-COUPLED 4TH
-ORDER FILTERS
In this section, the proposed mixed-coupling filter synthesis
method is applied to design two 4th
-order bandpass filters,
namely, Filter C and Filter D, as shown in Fig. 7(a) and 8(a).
Both filters are purposely design with the same in-band
specification, i.e., central frequency at 2.45 GHz, 20 dB
in-band return loss, and 7% fractional bandwidth. The
inter-resonator coupling M12 is chosen as inductive-dominant,
where1212
ˆˆ JK . Meanwhile, M34 is chosen as capacitive-
dominant, where3434
ˆˆ JK . The strip width of synchronously-
tuned resonators is set as 0.3 mm, thus its characteristic
impedance 𝑍𝑟 equals to 80.4 Ω. According to the 4th
-order
Chebyshev low-pass prototype [19] and (2),
0638.02920.19314.0
%7
21
12
gg
FBWM
(13)
and,
05.04
0638.04
| ˆˆ| 12
MJK (14)
05.0ˆˆ JK is chosen for the inductive-dominant M12 stage,
and 05.0ˆˆ KJ is chosen for the capacitive-dominant M34
stage.
Fig. 7(a) illustrates the final optimized Filter C with detailed
dimensions. Herein, the J12 realized with anti-parallel coupled
line with the dimensions of L=2.2 mm, S= 0.7mm and W=0.3
mm. At the filter central frequency 2.4 GHz, 12J equals to 0.06.
For the quasi-lumped K12 inverter with Lstub=1.1 mm, Wstub=0.6
mm and via diameter D=0.4 mm, 12K equals to 0.104.
Meanwhile, the dimensions for J34 inverter are derived as
L=2.1 mm, S=0.2 mm and W=0.3 mm, so we can extract34J
=0.14. The quasi-lumped inverter K34 has Lstub of 0.75 mm,
Wstub of 0.6 mm and metallic via diameter of 0.4 mm. The
extracted 34K =0.082 at 2.4 GHz. The mixed-couplings 0.044 (
12K -12J ) and 0.058 (
34J -34K ), slightly deviated from 0.05
calculated in (14). The coupling M23 in Filter C is implemented
with a modified parallel coupled line, its magnetic coupling
strength is much smaller than its electric counterpart, and it is
thus modeled as capacitive-only coupling J23.
Simulated, measured and ideal 4th
-order Chebyshev filter
frequency responses for Filter C are compared in Fig. 7(c).
In-band fullwave simulated frequency responses are well
matched with the ideal 4th
-order counterpart. The measured
in-band insertion and return losses are 2.4 and 10.6 dB,
respectively. The lower-side transmission zero fZ1 by
capacitive-dominant M34 locates at 2.16 and 2.13 GHz in
simulated and measured results, respectively. It slightly
deviates from the theoretical calculated 2.1 GHz. The
(a)
Input
Ji,1J1,2
K1,2
J2,3
J3,4
K3,4 J4,oOutput3 421
(b)
(c)
Fig. 7. (a) Physical layout of 4th-order Filter C, (b) mixed-coupling topology,
and (c) comparison between simulated and measured frequency responses.
TMTT-2015-01-0083
6
simulated 2nd
transmission zero fZ2 at 2.80 GHz is consistent
with measured 2.86 GHz. However, the 2nd
transmission zero
at 4.75 GHz due to inductive-dominant M12 stage disappears,
and it may be interpreted by the fact that the length ratio of
coupling path deviates from targeted k=0.43 in Fig. 7(a) M12
stage. In other word, the port 2 in Fig. 1(b) for M12 stage could
not be clearly identified due to the distributed nature of J23
stage. Thus, it somehow destroys the synthesized length ratios
for both M12 and M34 stage.
To solve this problem, an improved filter, named as Filter D
in Fig. 8(a) is proposed. The middle M23 stage is now realized
by a modified anti-parallel coupled line with an embedded
lumped inter-digital capacitor. After intensive analysis, there
are three main advantages from this capacitor. (a) The
anti-parallel coupled line without extra inter-digital capacitor
in Fig. 8(a) itself is a mixed coupling. Its electric and magnetic
strengths are comparable and cancel each other, resulting in
coupling strength too weak to realize a 7% fractional
bandwidth. Hence, the extra inter-digital capacitor enhances its
capacitive component inside M23, and improves the net
coupling strength. (b) The phase difference for the mixed
coupling stage M12 and M34 can now be flexibly controlled so
as to freely adjust transmission zeros positions using the
method discussed in Section III. (c) This capacitive-dominant
mixed coupling M23, modeled as J23 and K23, is also capable to
create one extra lower-side transmission zero fZ2, thereby
enhancing lower-side rejection skirt.
The fabricated Filter D and its corresponding coupling
topology are depicted in Fig. 8(a) and 8(b), respectively, where
the specified coupling parameters are given as 12J = 0.065 and
12K = 0.11 in the M12 stage, 34J =0.164 and
34K =0.114 in the
M34 stage. The inductive-dominant M12 creates two finite
transmission zeros at fZ3=2.75 GHz and fZ4=4.55 GHz in the
simulated frequency response. Meanwhile, the capacitive-
dominant M34 stage creates a lower-side transmission zero
fZ1=2.2 GHz. Moreover, the lumped inter-capacitor embedded
in M23 not only preserves the coupling path ratios for both M12
and M34 stages, but also brings an additional transmission zero
at fZ2= 2.26 GHz. After testing the fabricated filter, the
fullwave simulated, measured, and ideal 4th
-order Chebyshev
frequency responses for Filter D are plotted and compared in
Fig. 8(c). Good agreement between them has evidently verified
our proposed phase-dependent mixed-coupling formulation for
transmission zeros design.
Unfortunately, there are always a few uncontrollable
parasitic effects as commonly existed in the implementation
and measurement of planar bandpass filters, such as fabrication
tolerance in etching process, transition discontinuity between
the feeding line and SMA connector, inaccurate value of
permittivity listed in the datasheet, nonzero conductor
thickness and so on. As a result, the measured 10 dB return
losses of 4th
-order filters, as shown in Fig. 6(c) to Fig. 8 (c), do
not perfectly achieve the targeted 20 dB in design.
V. CONCLUSION
In this paper, a general synthesis method for the inline
mixed-coupled quasi-elliptic bandpass filters is proposed. The
mixed inter-resonator coupling is modeled in such a way that
both magnitudes and phase differences of constituted
capacitive and inductive couplings are made use of in finite
transmission zeros synthesis. Two 2nd
-order filters with
identical inductive-dominant coupling and different length
ratios are analytically synthesized. Next, two 4th
-order
quasi-elliptic bandpass filters with both inline capacitive- and
inductive-dominant mixed couplings are presented, designed
and fabricated. The multiple finite transmission zeros in
measured frequency responses of the fabricated filters have
well justified our proposed mixed-coupling formulation and
transmission zeros synthesis.
ACKNOWLEDGMENT
The authors are grateful to Mr. Li Yang and Dr. Dong Chen
from University of Macau for their kind help on circuit
fabrication and measurements.
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(a)
Ji,1J1,2
K1,2
J3,4
K3,4
J4,oJ2,3
K2,3
Input Output1 32 4
(b)
(c)
Fig. 8. (a) Physical layout of 4th-order Filter D, (b) mixed-coupling topology,
and (c) comparison between simulated and measured frequency responses.
TMTT-2015-01-0083
7
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Songbai Zhang (S’11–M’14) received the B.
Eng. and Ph.D degrees in Electrical and
Electronics Engineering from Nanyang
Technological University (NTU), Singapore, in
2010 and 2014, respectively. From Jul. 2013 to Jan. 2015, he was with the
Institute of Microelectronics, Agency for
Science, Technology and Research (A*Star), Singapore. His research includes RF/microwave
passive devices: filters and antennas, advanced
3D-IC package and RFICs. Since Feb. 2015, he is with Skyworks Solutions as a Senior RFIC
Design Engineer, involved in mobile transceiver
frontend power amplifier and antenna switch module design. Dr. Zhang was a recipient of the Ministry of Education Scholarship,
Singapore, from 2006 to 2010, and the NTU Research Scholarship from 2010
to 2014. He was the best student paper recipient of the 2013 International Wireless Symposium, Beijing. He also served on as a reviewer for IEEE
TRANSACTION ON MICROWAVE THEORY AND TECHNIQUES and IEEE
MICROWAVE AND WIRELESS COMPONENTS LETTERS.
Lei Zhu (S’91–M’93–SM’00–F’12) received
the B.Eng. and M.Eng. degrees in radio engineering from the Nanjing Institute of
Technology (now Southeast University),
Nanjing, China, in 1985 and 1988, respectively, and the Ph.D. degree in electronic engineering
from the University of Electro-
Communications, Tokyo, Japan, in 1993. From 1993 to 1996, he was a Research
Engineer with Matsushita-Kotobuki Electronics
Industries Ltd., Tokyo, Japan. From 1996 to 2000, he was a Research Fellow with the École Polytechnique de Montréal,
Montréal, QC, Canada. From 2000 to 2013, he was an Associate Professor
with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore. Since August 2013, he has been a
Professor with the Faculty of Science and Technology, University of Macau,
Macau, China. Since September 2014, he has been serving as the Head of Department of Electrical and Computer Engineering, University of Macau. His
research interests include microwave circuits, guided-wave periodic structures,
antennas, and computational electromagnetic techniques. Dr. Zhu was the Associate Editors for the IEEE TRANSACTION ON
MICROWAVE THEORY AND TECHNIQUES (2010-2013) and the IEEE
MICROWAVE AND WIRELESS COMPONENTS LETTERS (2006-2012). He served as a General Chair of the 2008 IEEE MTT-S International Microwave
Workshop Series on the Art of Miniaturizing RF and Microwave Passive
Components, Chengdu, China, and a Technical Program Committee Co-Chair of the 2009 Asia–Pacific Microwave Conference, Singapore. He has been
serving as the members of IEEE MTT-S Fellow Evaluation Committee and
IEEE AP-S Fellows Committee since 2013 and 2015, respectively. He was the
recipient of the 1997 Asia–Pacific Microwave Prize Award, the 1996 Silver
Award of Excellent Invention from Matsushita-Kotobuki Electronics Industries Ltd., and the 1993 First-Order Achievement Award in Science and
Technology from the National Education Committee, China.
Roshan Weerasekera (S’01–M’04–SM’12)
received the B.Sc. degree from the University of
Peradeniya, Peradeniya, Sri Lanka, in 1998 and the M.Sc. and Ph.D. degrees in electronic system
design from the Royal Institute of Technology,
Stockholm, Sweden, in 2002 and 2008, respectively.
From October 2001 to December 2003, he
was a Lecturer in the Department of Electrical and Electronic Engineering, University of
Peradeniya, and from 2008 to 2011, he was a
Research Associate with Lancaster University, U.K., within a 3-D Flash memory integration
project (ELITE) under a European Union FP7 research grant. Currently, he is
with the Institute of Microelectronics (IME), Agency for Science, Technology and Research (A*STAR), Singapore. His research interests include the
2D/2.5D/3D heterogeneous system design for applications such as data-center
servers, portable computing platforms; and integrated bio-sensor platforms for point-of-care systems.