LINEAR TECHNOLOGYLINEAR TECHNOLOGY · Linear Technology Magazine • May 2002 3 LTC1733, continued...

MAY 2002 VOLUME XII NUMBER 2

LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLOGY

, LTC, LT, Burst Mode, OPTI-LOOP and Over-The-Top are registered trademarks of Linear Technology Corporation.Adaptive Power, C-Load, DirectSense, FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD, MultimodeDimming, No Latency ∆Σ, No RSENSE, Operational Filter, PolyPhase, PowerSOT, SoftSpan, SwitcherCAD, ThinSOT andUltraFast are trademarks of Linear Technology Corporation. Other product names may be trademarks of the companiesthat manufacture the products.

IN THIS ISSUE…COVER ARTICLE

LTC1733: Thermal RegulationMaximizes Lithium-Ion BatteryCharging Rate Without Risk ofOverheating .................................. 1Trevor Barcelo

Issue Highlights ............................ 2

LTC® in the News ........................... 2

DESIGN FEATURES

LT®3420 Charges PhotoflashCapacitors Quickly and EfficientlyWhile Using Minimal Board Space..................................................... 5

Albert Wu

Small 1.25A Step-Down RegulatorSwitches at 4MHz for Space-SensitiveApplications .................................. 9Damon Lee

Dual DC/DC Controller Brings 2-PhaseBenefits to Low Input VoltageApplications ................................ 12Jason Leonard

ThinSOT™ RF Power Controllers SaveCritical Board Space and Power inPortable RF Products ................... 15Ted Henderson and Shuley Nakamura

DESIGN IDEAS.............................................. 21–35

(complete list on page 21)

New Device Cameos ..................... 36

Design Tools ................................ 39

Sales Offices ............................... 40

continued on page 3

IntroductionLinear battery chargers are typicallysmaller, simpler and less expensivethan switcher-based solutions, butthey have one major disadvantage:excessive power dissipation. Whenthe input voltage is high and thebattery voltage is low (dischargedbattery) a linear charger could gener-ate enough heat to damage itself orother components. Typically, suchconditions are temporary—as thebattery voltage rises with its charge—but it is worst case situations thatone must account for when determin-ing the maximum allowable valuesfor charge current and IC tempera-ture. To solve this problem, theLTC1733 employs internal thermal

LTC1733: ThermalRegulation MaximizesLithium-Ion BatteryCharging Rate WithoutRisk of Overheating

by Trevor Barcelofeedback to regulate the charge cur-rent and limit the die temperature.This feature translates to faster chargetimes, because a designer can pro-gram a high charge current (tominimize charging time) without therisk of damaging the LTC1733 or anyother components. The need for ther-mal over-design is also eliminated. Tofurther improve heat transfer, theLTC1733 is housed in a thermallyenhanced 10-pin MSOP package. Forsimplicity, the LTC1733 provides acomplete lithium-ion charger solu-tion requiring only three externalcomponents, as shown in Figure 1.

An internal power MOSFET allowscharge current to be programmed up

BAT

PROGTIMER

GND NTC

VCC

VIN = 5V

1.5k1%

4.7µF

0.1µF

IBAT = 1A

1733TA01

LTC1733 1-CELLLi-IonBATTERY*

*AN OUTPUT CAPACITOR MAY BE REQUIRED DEPENDING ON BATTERY LEAD LENGTH

Figure 1. Standalone Li-Ion battery charger

Linear Technology Magazine • May 20022

EDITOR’S PAGE

For further information on anyof the devices mentioned in thisissue of Linear Technology, usethe reader service card or callthe LTC literature servicenumber:

1-800-4-LINEAR

Ask for the pertinent data sheetsand Application Notes.

Issue Highlights LTC in the News…On April 16, Linear Technology

Corporation announced its finan-cial results for the 3rd quarter offiscal year 2002. According to Rob-ert H. Swanson, Chairman of theBoard and CEO, “For the first timein several quarters, all of the criti-cal financial trends showed healthyimprovement. Sales and profitsgrew sequentially 7% and 12% re-spectively over the previous quarter.Bookings, which exceeded sales,grew in all major geographical andmajor end markets. Even at thesereduced sales levels from last year,we continue to be strongly profit-able with a 40% after tax return onsales. In January, we discontinuedproduction in our oldest wafer fab-rication plant. The associated costshad been previously provided forin past financial statements, andtherefore, no special one-timecharges were required.

Looking forward, we have seen abroad based increase in our book-ings activity throughout thequarter. However, our backlog,while improving, is still low, andglobal economic and political con-ditions continue to be tenuous.Therefore, confidently and accu-rately forecasting future financialresults remains difficult. However,based on analysis of the data avail-able to us, we believe excessinventory of our product at cus-tomers continues to be worked offand we expect bookings to con-tinue to improve. Consequently,we forecast sales and profits togrow sequentially in the 8% to 10%range from the quarter just com-pleted.”

The Company reported net salesof $130,155,000 and net income of$51,480,000 for the quarter endedMarch 31, 2002. Diluted earningswere $0.16 per share.

Our cover article introduces theLTC1733 battery charger, which em-ploys internal thermal feedback toregulate the charge current and limitthe die temperature. This featuretranslates to faster charge times, be-cause a designer can program a highcharge current (to minimize chargingtime) without the risk of damagingthe LTC1733 or any other compo-nents. The need for thermalover-design is also eliminated. To fur-ther improve heat transfer, theLTC1733 is housed in a thermallyenhanced 10-pin MSOP package. Forsimplicity, the LTC1733 provides acomplete lithium-ion charger solu-tion requiring only three externalcomponents.

The remainder of the Design Fea-tures section presents a variety ofpower products:

The LT3420 is a power IC that isdesigned for charging large-valuedcapacitors to high voltages, such asthose used for the strobe flashes ofdigital and film cameras. Using theLT3420, only a few external compo-nents are necessary to create acomplete solution, which saves valu-able board space in ever shrinkingcamera designs.

The LTC3411 DC/DC converterprovides features that shrink totalsolution size enough to fit into thelatest portable electronics. Its capableof switching frequencies as high as4MHz, allowing the use of smaller

and less costly capacitors and induc-tors to complete the circuit. It alsosaves space by placing the switcherand MOSFETs in a small monolithicpackage.

The LTC3701 is another space sav-ing power product. It is an efficient,low input voltage, dual DC/DC con-troller that fits into tight spaces. Ituses 2-phase switching techniquesto reduce required input capacitance(saving space and cost) and increaseefficiency. The versatile LTC3701 ac-cepts a wide range of input voltages,from 2.5V to 9.8V, making it usefulfor single lithium-ion cell and manymulticell systems. It can provide out-put voltages as low as 0.8V and outputcurrents as high as 5A.

The LTC4400-1 and LTC4401-1provide RF power controller solutionsfor the latest cellular telephones. Theyfeature very small footprints, lowpower consumption and wide fre-quency ranges while minimizingadjacent channel interference by care-fully controlling RF power profiles.The LTC4400-1 and LTC4401-1 areboth available in a low profile 6-pinThinSOT package, and require fewexternal parts. For example, whenused with a directional coupler, onlytwo resistors and two capacitors arerequired. Both devices require mini-mal power to operate, typically 1mAwhen enabled and 10µA when inshutdown.

Starting on page 21 are nine newDesign Ideas covering a variety ofapplications, from a lightweight por-table altimeter, to a way to create aVCO from the LTC6900 precisionoscillator, to a simple way to createtwo lowpass filters out of a singlefilter IC. See page 21 for a completelist of the Design Idea articles.

At the back are seven New DeviceCameos. See www.linear.com for com-plete device specifications and moreapplications information.

Authors can be contactedat (408) 432-1900

DESIGN FEATURES

Linear Technology Magazine • May 2002 3

LTC1733, continued from page 1to 1.5A, with 7% accuracy, to ensurea fast and complete charge. The inter-nal MOSFET also eliminates the needfor an external current sense resistoror blocking diode. The final float volt-age is pin selectable to either 4.1V or4.2V with 1% accuracy to preventdangerous overcharging or reducedbattery capacity due to undercharg-ing.

Following battery manufacturers’guidelines, the LTC1733 includes aprogrammable charge terminationtimer and thermistor input for tem-perature qualified charging. Statusoutputs include C/10 charge detec-tion to indicate a near end-of-chargecondition, wall adapter present de-tection to determine whether chargingmay proceed or not, charge currentmonitoring for gas gauging, and faultdetection for identifying bad cells.Low battery charge conditioning(trickle charging) safely charges anover-discharged cell, and automaticrecharge ensures that the battery isalways fully charged. To conservebattery power, the LTC1733 batterydrain current drops to less than 5µAwhen a wall adapter is not present orwhen the part is shutdown.

Charging a BatteryTo charge a single cell Li-ion battery,the user must apply an input voltage(typically, a wall adapter) of at least4.5V to the VCC pin. The ACPR pin willsubsequently pull low to indicate thatthe input voltage condition has beenmet. Furthermore, a 1% resistor mustbe connected from PROG to GND toprogram the nominal charge currentto 1500V/RPROG. The CHRG pin willthen pull low to indicate that a chargecycle has commenced. A capacitorconnected between the TIMER pinand GND will program the chargetermination time to 3 hours per 100nF.

If the BAT pin voltage is below2.48V at the beginning of a chargecycle then the charge current will beone-tenth of the programmed valuein order to safely bring the cell voltagehigh enough to allow full charge cur-rent. If the cell is damaged, and thevoltage does not rise above 2.48V

within one-quarter of the programmedtermination time, the charge cyclewill terminate, and the FAULT statusoutput will latch low indicating a badcell. All three of these status outputpins, ACPR, CHRG and FAULT, haveenough current sinking capability tolight an LED.

Once the battery voltage rises above2.48V (which typically occurs soonafter the start of a charge cycle), theLTC1733 will provide a constant cur-rent to the battery as programmed byRPROG. The LTC1733 will remain inconstant-current mode until the BATpin voltage approaches the selectedfinal float voltage (4.1V for SEL = 0Vand 4.2V for SEL = VCC). At this pointthe part enters constant-voltage mode.

In constant-voltage mode, theLTC1733 begins to decrease thecharge current to maintain a con-stant voltage at the BAT pin rather

than a constant current out of theBAT pin. When the current drops to10% of the full-scale programmedcharge current, an internal compara-tor latches off the strong pull-down atthe CHRG pin and connects a weakcurrent source (about 25µA) to groundto indicate a near end-of-charge (C/10) condition.

Unlike battery chargers that ter-minate when the current reaches C/10, the LTC1733 continues to chargethe battery after the C/10 point, aslong as the termination time has notelapsed, to ensure that the battery isfully charged. Terminating chargingat C/10 can leave a battery charged toonly 90% to 95% capacity, while charg-ing past C/10 and terminating basedon time can charge a battery to 100%capacity. Upon termination, theCHRG pin assumes a high impedancestate.

Recharging a BatteryThe LTC1733 has the ability to re-charge a battery assuming that thebattery voltage has been chargedabove 3.95V (SEL = 0V) or 4.05V (SEL= VCC) during the initial charge cycle.Once above these thresholds, a newcharge cycle begins if the battery volt-age drops below 3.9V (SEL = 0V) or4.0V (SEL = VCC) due to either a loadon the battery, or the self-dischargecurrent of the battery. The rechargecircuit integrates the BAT pin voltagefor a few milliseconds to prevent tran-sients from restarting the charge cycle.This feature ensures that the batteryremains charged even if left connectedto the powered charger for very longperiods of time.

Thermal RegulationAn additional key feature of theLTC1733 is the internal thermal regu-lation loop. If high power operationand/or high ambient temperatureconditions cause the junction tem-perature of the LTC1733 to approach105°C, the charge current is auto-matically reduced to maintain thejunction temperature at roughly105°C (board temperatures typicallyremain below about 85°C). This iscalled constant-temperature mode.

–

+

–

+

–

+

1733 F02

RNTC10k

RHOT1%

7/8 VCC

1/2 VCC

3/160 VCC

TOOCOLD

TOOHOT

DISABLENTC

LTC1733

VCC

NTC

Figure 3. Temperature qualification circuitry

Figure 2. Full featured single cellLi-Ion charger


DESIGN FEATURES

This feature allows the user to pro-gram a charge current based on typicaloperating conditions and eliminatesthe need for the complicated thermalover-design necessary in many linearcharger applications. Worst-case con-ditions are automatically taken careof by the LTC1733. In addition toprotecting the LTC1733, this featureeliminates “hot spots” on the board,thereby protecting surrounding com-ponents. The thermal shutdownfeatures of other battery chargers sim-ply turn off the charger at very hightemperatures (typically, in excess of130°C). This junction-temperature-based type of shutdown allows boththe battery charger and the surround-ing board to get extremely hot, soeven though the shutdown “protec-tion” exists, the application must bepainstakingly designed to avoidreaching the thermal shutdown tem-perature under all scenarios. TheLTC1733 simplifies thermal designby automatically balancing chargecurrent, power dissipation and oper-ating temperature.

To further improve the thermalperformance of the LTC1733, it ispackaged in a 10-pin thermallyenhanced MSOP package. The appli-cation board pictured in Figure 2occupies just 76mm2 of board spaceand can dissipate over 2W of power atroom temperature. That equates to amaximum charge current of about1.5A, with a 5V input supply. Thisassumes that a Li-ion battery spendsmost of its time at 3.7V during charge.

In fact, this is a conservative assump-tion, since a typical Li-ion battery willrise above 3.8V within the first fewminutes of charging. The powerfulthermal features of the LTC1733 andthe 7% accuracy of the programmedcharge current allow very fast andaccurate charging of single cell Li-ionbatteries.

PROG Current MonitorFor gas gauging applications, thePROG pin provides very accurate in-formation regarding the currentflowing out of the BAT pin. The rela-tionship is given by:

IBAT = (VPROG/RPROG) • 1000During constant-current mode, the

PROG pin voltage is always 1.5V,indicating that the programmedcharge current is flowing out of theBAT pin. In constant-temperature orconstant-voltage mode, the BAT pincurrent is reduced and can be deter-mined by measuring the PROG pinvoltage and applying the above for-mula. The PROG pin, along with thethree open-drain status outputs(ACPR, CHRG, and FAULT), informthe user of exactly what the LTC1733is doing at all times.

NTC ThermistorIn addition to the programmable timerand low battery charge qualification,the LTC1733 adds temperature quali-fied charging to the list of batterymanufacturer recommended safetyfeatures. The battery temperature is

measured by placing a negative tem-perature coefficient (NTC) thermistorclose to the battery pack. Using thecircuitry shown in Figure 3, theLTC1733 can temporarily suspendthe internal timer and stop chargingwhen the battery temperature fallsbelow 0°C or rises above 50°C. Toperform this function, RHOT should bechosen to be the value of the selectedNTC thermistor at 50°C. This willensure that the internal comparator’strip point of 1/2VCC corresponds toan NTC temperature of 50°C. Fur-thermore, the selected NTC thermistorshould have a value at 0°C that is asclose to seven times the value at 50°Cas possible. A 7:1 cold to hot NTCratio ensures that the internalcomparator’s trip point of 7/8VCCcorresponds to an NTC temperatureof 0°C. The hot and cold comparatorseach have approximately 2°C of hys-teresis to prevent oscillation aboutthe trip point. In addition, the NTCfunction can be disabled without anyexternal components by simplygrounding the NTC pin.

ConclusionThe LTC1733 is a full-featured,standalone Li-ion battery charger. Inits simplest form, the LTC1733 onlyrequires three external componentsand can safely and accurately chargehigh-capacity batteries very quicklywith up to 1.5A of charge current. AnNTC thermistor and a few LEDs canbe added to take advantage of thesafety and status features.

DESIGN FEATURES


IntroductionThe LT3420 is a power IC, designedprimarily for charging large-valuedcapacitors to high voltages, such asthose used for the strobe flashes ofdigital and film cameras. These ca-pacitors are generally referred to asphotoflash or strobe capacitors andrange from values of a hundred mi-crofarads to a millifarad, with targetoutput voltages above 300V. Thephotoflash capacitor is used to storea large amount of energy, which canbe released nearly instantaneously topower a xenon bulb, providing thelight necessary for flash photogra-phy. Traditional solutions for chargingthe photoflash capacitor, such as theself-oscillating type, are extremelyinefficient. More modern techniquesuse numerous discrete devices toimplement a flyback converter butrequire a large board area and sufferfrom high peak currents, reducingbattery life. The LT3420 incorporatesa low resistance integrated switchand utilizes a new patent-pendingcontrol technique to solve this diffi-cult high voltage power problem. Usingthe LT3420, only a few external com-ponents are necessary to create acomplete solution, which saves valu-able board space in ever shrinkingcamera designs. Efficiency of the

LT3420 is high, typically greater than75%, while the peak current of thepart is well controlled, important fea-tures for increasing battery life.

OverviewFigure 1a shows a photoflash appli-cation for the LT3420. To generatethe high output voltage required, theLT3420 is designed to operate in aflyback switching regulator topology.The LT3420 uses an adaptive on-time/off-time control scheme resulting inexcellent efficiency and precise con-trol of switching currents. The LT3420can charge a 220µF capacitor from50V to 320V in 3.5s from a 5V input,as shown in Figure 1b. Charge timedecreases with higher VIN, as shownin Figure 1c. 50V is used as thestarting point in calculating chargetime since the xenon bulb will selfexinguish at this voltage, halting any

further voltage drop on the photoflashcapacitor.

In Figure 1a, the circuitry to theright of C4 shows a typical way togenerate the light pulse once thephotoflash capacitor is charged. Whenthe SCR is fired, the flying lead placednext to the xenon bulb reaches manykilovolts in potential. This ionizes the

LT3420 Charges Photoflash CapacitorsQuickly and Efficiently While UsingMinimal Board Space by Albert Wu

3420 F01

+VCC

CHARGEDONE

SEC

RREF

LT3420

VBAT RFB SW

CT GND

C24.7µF

C30.1µF

C14.7µF

VBAT1.8V TO 10V

VCC2.5V TO 10V

CHARGEDONE

R151.1k

T11:12

320V

3,4

5,6 8

1

R22k

D1

FLASH

FLYINGLEAD

C1, C2: 4.7µF, X5R or X7R, 10VC4: RUBYCON 220µF PHOTOFLASH CAPACITOR (858) 496-8990T1: TDK SRW10EPC-U01H003 FLYBACK TRANSFORMER (408) 392-1400D1: GENERAL SEMICONDUCTOR GSD2004S SOT-23 (516) 847-3000

DUAL DIODE. DIODES CONNECTED IN SERIES

DANGER HIGH VOLTAGEOPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY

C4220µF330V

PHOTOFLASHCAPACITOR

BOLD LINES INDICATEHIGH CURRENT PATHS

Figure 1a. 320V photoflash capacitor charging circuit

VOUT50V/DIV

VCHARGE5V/DIV

1s/DIV 3420 F01bVBAT = 5V

Figure 1b. Charging waveform

VBAT (V)2

TIM

E (s

)

103420 G03

4 6 8

10

8

6

4

2

0

VOUT CHARGED FROM50V TO 320V

COUT = 220µF

COUT = 100µF

Figure 1c. Charge time


DESIGN FEATURES

gas inside the bulb forming a lowimpedance path across the bulb. Theenergy stored in the photoflash ca-pacitor quickly flows through theXenon bulb, producing a burst oflight. It is important to implement theground routing shown in Figure 1a,because during a flash, hundreds ofamps can flow in the traces indicatedby bold lines. Improper ground rout-ing can result in erratic behavior ofthe circuit.

Figure 2 shows a simplified blockdiagram of the LT3420. At any giveninstant, the Master Latch determineswhich one of two modes the LT3420 isin: “Power Delivery” or “Refresh.” InPower Delivery Mode, the circuitryenclosed by the smaller dashed box isenabled, providing power to chargephotoflash capacitor C4. The outputvoltage is monitored via the flybackpulse on the primary of the trans-former. Since no output voltage divideris needed, a significant source of powerloss is removed. In fact, the only DCloading on the output capacitor isdue to inherent self-leakage of the

capacitor and minuscule leakage fromthe rectifying diode. This results inthe photoflash capacitor being able toretain most of its energy when theLT3420 is in shutdown.

Once the target output voltage isreached, the power delivery mode isterminated and the part enters therefresh mode. In refresh mode, thepower delivery block is disabled, re-ducing quiescent current, while therefresh timer is enabled. The refreshtimer simply generates a user pro-grammable delay, after which the partreenters the power delivery mode.Once in the power delivery mode, the

3420 BD

–

+

–

+

–

+

+ –

RS Q

ONE-SHOT

1VREFERENCE

5

8

9

2 3 1 6

7

MASTERLATCH

Q Q

S R

DONE VBAT

VBAT

RFB SW

GND

RREF

SEC

VCC VCC

VOUT

CHARGE

CHIPENABLE

CT

BLOCKENABLE

Q5 Q3

C3

C2

C1

Q4

Q2

Q1D3

A3 A2

A1

REFRESHTIMER

ONE-SHOT

ENABLE

10

4

+ –

LT3420

R2

R1

PRIMARY SECONDARY

+

0.014Ω

0.25Ω

20mV

10mV

DRIVER

POWER DELIVERY BLOCK

T11:12

D1

C4PHOTOFLASH

CAPACITOR

Figure 2. Simplified block diagram of the LT3420

VOUT100V/DIV

MODE

IIN1A/DIV

VCT1V/DIV

1s/DIV 3420 F03

SHUTDOWN CHARGING REFRESH

Figure 3. The three operating modesof the LT3420: shutdown, charging,and refresh of the photoflashcapacitor

ISW1A/DIV

VSW20V/DIV

ISEC200mA/DIV

2µs/DIV 3420 F04a 2µs/DIV 3420 F04b

ISW1A/DIV

VSW20V/DIV

ISEC200mA/DIV

Figure 4b. Switching waveforms withVOUT = 300V, VCC=VBAT = 3.3V

Figure 4a. Switching waveforms withVOUT = 100V, VCC=VBAT = 3.3V


DESIGN FEATURES

LT3420 will again provide power tothe output until the target voltage isreached. Figure 3 is an oscillogramshowing both the initial charging ofthe photoflash capacitor and the sub-sequent refresh action. The upperwaveform is the output voltage. Themiddle waveform is the voltage on theCT pin. The lower waveform showsthe input current. The mode of thepart is indicated below the photo.

The user can defeat the refreshtimer and force the part into powerdelivery mode by toggling the CHARGEpin high then low, then high again.The low-to-high transition on theCHARGE pin fires a one-shot thatsets the master latch, putting thepart in power delivery mode. BringingCHARGE low puts the part in shut-down. The refresh timer can beprogrammed to wait indefinitely bysimply grounding the CT pin. In thisconfiguration, the LT3420 will onlyreenter the power delivery mode bytoggling the CHARGE pin.

In power delivery mode, the LT3420operates by adaptively controlling theswitch on-time and off-time. Theswitch on-time is controlled so thatthe peak primary current is 1.4A (Typi-cal). The switch off-time is controlledso the minimum secondary current is40mA (Typical). With this type of con-trol scheme, the part always operatesin the CCM (Continuous ConductionMode), resulting in rapid charging ofthe output capacitor. A side benefit ofthis scheme is that the part can sur-vive a short circuit on the outputindefinitely. Figure 4a and 4b show

the relevant currents during the powerdelivery mode when VOUT is 100V and300V respectively. Notice how the on-time and off-time are automaticallyadjusted to keep the peak current inthe primary and secondary of thetransformer constant as VOUT in-creases.

Measuring EfficiencyMeasuring the efficiency of a circuitdesigned to charge large capacitiveloads is a difficult issue, particularlywith photoflash capacitors. The ideal

way to measure the efficiency of acapacitor charging circuit would be tofind the energy delivered to the out-put capacitor (0.5 • C • V2) and divideit by the total input energy. Thismethod does not work well here be-cause photoflash capacitors are farfrom ideal. Among other things, theyhave relatively high leakage currents,large amounts of dielectric absorp-tion, and significant voltagecoefficients. A much more accurate,and easier, method is to measure theefficiency as a function of the output

3420 TA01

+

VCC

CHARGEDONE

SEC

RREF

LT3420

VBAT RFB SW

CT GND

C24.7µF

C30.1µF

C14.7µF

VBAT1.8V TO 10V

VCC2.5V TO 10V

R152.3k

T11:12

320V

2

3 4

1

R22k

D1

C1, C2, C4, C5, C6, C7: 4.7µF, X5R or X7R, 10VT1-T3: PULSE PA0367 FLYBACK TRANSFORMER (619) 674-8100D1-D3: GENERAL SEMICONDUCTOR GSD2004S SOT-23 (516) 847-3000

DUAL DIODE. DIODES CONNECTED IN SERIESQ1: 2N3904 OR EQUIVALENT

* CAN CHARGE ANY SIZE PHOTOFLASH CAPACITOR** USE AS MANY SLAVE CHARGERS AS NEEDED.

DANGER HIGH VOLTAGEOPERATION BY HIGH VOLTAGE TRAINED PERSONEL ONLY

650µF*350VPHOTOFLASHCAPACITOR

VCC

CHARGEDONE

SEC

RREF

LT3420

VBAT RFB SW

CT GND

C54.7µF

C44.7µF

VBAT

T21:12

2

3 4

1

D2

SLAVE**CHARGER

MASTERCHARGER

SLAVE**CHARGER

VCC

CHARGEDONE

SEC

RREF

LT3420

VBAT RFB SW

CT GND

C74.7µF

C64.7µF

VBAT

T31:12

2

3 4

1

D3

R4100k

R3100k

VCC

VCC

Q12N3904

CHARGE

Figure 6. This professional grade charger uses multiple circuitsin parallel to quickly charge large photoflash capacitors.

EFFI

CIEN

CY (%

)

3420 G10

90

80

70

60

50

40

VOUT (V)100 200 30050 150 350250

VCC = VBAT = VIN

VIN = 3.3V

VIN = 5V

Figure 5. Efficiency for the circuitin Figure 1


DESIGN FEATURES

VOUT50V/DIV

CHARGENO

CHARGE0.5s/DIV 3420 F05

5V/DIV

VCHARGE

Figure 7. Halting the charge cycle at any time

Figure 9. Input current as dutycycle is varied

INPU

T CU

RREN

T (m

A)

DUTY CYCLE (%)9010

800

030 50 70

400

200

600

A1

A3A2

TOLT3420CIRCUIT

CHARGEDONE

ON

1kHz PWMSIGNAL

Figure 8. Simple logic for adjustable inputcurrent

voltage. In place of the photoflashcapacitor, use a smaller, high qualitycapacitor, reducing errors associatedwith the non-ideal photoflash capaci-tor. Using an adjustable load, theoutput voltage can be set anywherebetween ground and the maximumoutput voltage. The efficiency is mea-sured as the output power (VOUT •IOUT) divided by the input power (VIN •IIN). Figure 5 shows the efficiency forthe circuit in Figure 1, which wasmeasured using this method. Thismethod also provides a good means tocompare various charging circuitssince it removes the variability of thephotoflash capacitor from the mea-surement. The total efficiency of thecircuit, charging an ideal capacitor,would be the time average of the givenefficiency curve, over time as VOUTchanges.

Standard TransformersLinear Technology Corporation hasworked with several transformermanufacturers (including TDK, Pulseand Sumida) to provide transformerdesigns optimized for the LT3420 thatare suitable for most applications.Please consult with the transformermanufacturer for detailed informa-tion. If you wish to design your owntransformer, the LT3420 data sheetcontains a section on relevant issues.

Professional PhotoflashChargerFigure 6 shows a professional gradecharger designed to charge large(>500µ F) photoflash capacitorsquickly and efficiently. Here, multipleLT3420 circuits can be used in paral-

lel. The upper most circuit in thefigure is the master charger. It oper-ates as if it were the only charger inthe circuit. The DONE signal fromthis charger is inverted by Q1 anddrives the CHARGE pin of all theother slave chargers. Notice thatgrounding the RREF and CT pinsdisables the control circuitry of theSlave chargers. The charging time fora given capacitor is inversely propor-tional to the number of chargers used.Three chargers in parallel takes athird of the charging time as a singlecharger applied to the same photoflashcapacitor. This circuit can charge a650µF capacitor from 50V to 320V in3.5s from a 5V input.

Interfacing to aMicrocontrollerThe LT3420 can be easily interfacedto a microcontroller. The CHARGEand DONE pins are the control andmode indicator pins, respectively, forthe part. By utilizing these pins, theLT3420 can be selectively disabledand enabled at any time. The

microcontroller can have full controlof the LT3420. Figure 7 shows theLT3420 circuit being selectively dis-abled when the CHARGE pin is drivenlow midway through the charge cycle.This might be necessary during asensitive operation in a digital cam-era. Once the CHARGE pin is returnedto the high state, the charging contin-ues from where it left off.

Adjustable Input CurrentWith many types of modern batteries,the maximum allowable current thatcan be drawn from the battery islimited. This is generally accomplishedby active circuitry or a polyfuse. Dif-ferent parts of a digital camera mayrequire high currents during certainphases of operation and very little atother times. A photoflash chargingcircuit should be able to adapt tothese varying currents by drawingmore current when the rest of thecamera is drawing less, and vice-versa. This helps to reduce the chargetime of the photoflash capacitor, whileavoiding the risk of drawing too muchcurrent from the battery. The inputcurrent to the LT3420 circuit can beadjusted by driving the CHARGE pinwith a PWM (Pulse Width Modulation)signal. The microprocessor can ad-just the duty cycle of the PWM signalto achieve the desired level of inputcurrent. Many schemes exist toachieve this function. Once the targetoutput voltage is reached, the PWMsignal should be halted to avoid over-charging the photoflash capacitor,since the signal at the CHARGE pinoverrides the refresh timer.

A simple method to achieve adjust-able input current is shown in Figure8. The PWM signal has a frequency of1kHz. When ON is logic high, thecircuit is enabled and the CHARGEpin is driven by the PWM signal. When

continued on page 11


DESIGN FEATURES

IntroductionCell phones, pagers, PDAs and otherportable devices are shrinking, andas they shrink, the demand for smallercomponents grows. The ubiquitousswitching regulator, which solves theproblem of creating a constant volt-age from inconstant batteries, is notexempt from the demand to becomesmaller. One way to shrink regulatorcircuitry is to increase the switchingfrequency of the regulator, allowingthe use of smaller and less costlycapacitors and inductors to completethe circuit. Another way is to shrinkthe switcher itself by putting theswitcher and MOSFETs in a smallmonolithic package. The LTC3411DC/DC converter does both.

The LTC3411 is a 10-lead MSOP,synchronous, step-down, currentmode, DC/DC converter, intended formedium power applications. It oper-ates within a 2.5V to 5.5V input voltagerange and switches at up to 4MHz,making it possible to use tiny capaci-tors and inductors that are under2mm in height. By using the LTC3411in a small MS10 package, a completeDC/DC converter can consume lessthan 0.3 square inches of board realestate, as shown in Figure 1.

The output of the LTC3411 is ad-justable from 0.8V to 5V. For

battery-powered applications thathave input voltages above and belowthe output, the LTC3411 can also beused in a single inductor, positiveBuck-Boost converter configuration.A built-in 0.11Ω switch allows up to1.25A of output current at high effi-ciency. OPTI-LOOP® compensationallows the transient response to beoptimized over a wide range of loadsand output capacitors.

Efficiency takes on grave impor-tance in battery-powered applications,and the LTC3411 keeps efficiencyhigh. Automatic, power saving BurstMode® operation reduces gate chargelosses at low load currents. With noload, the converter draws only 62µA,and in shutdown, it draws less than

1µA, making it ideal for low currentapplications.

The LTC3411 uses a current mode,constant frequency architecture thatbenefits noise sensitive applications.Burst Mode operation is an efficientsolution for low load current ap-plications, but sometimes noisesuppression takes on more impor-tance than efficiency, especially intelecommunication devices. To reducenoise problems, the LTC3411 pro-vides a pulse-skipping mode and aforced-continuous mode. Thesemodes decrease the ripple noise andimprove noise filterability. Althoughnot as efficient as Burst Modeoperation at low load currents, pulse-skipping mode and forced continuousmode can still provide high efficiencyfor moderate loads (see Figure 3). Indropout, the internal P-channel MOS-FET switch is turned on continuously,thereby maximizing the usable bat-tery life.

A High Efficiency 2.5V Step-Down DC/DC Converter withall Ceramic CapacitorsThe low cost and low ESR of ceramiccapacitors make them a very attrac-

Small 1.25A Step-Down RegulatorSwitches at 4MHz for Space-SensitiveApplications by Damon Lee

Figure 1. A complete DC/DC converter cantake less than 0.3in2 of board space.

LOAD CURRENT (mA)

EFFI

CIEN

CY (%

)

100

95

90

85

80

75

70

65

601 100 1000 10000

3411 G04

10

VIN = 3.3VVOUT = 2.5VCIRCUIT OF FIGURE 6

BURST MODEOPERATION

PULSE SKIP FORCE CONTINUOUS

Figure 3. Efficiencies for the circuitshown in Figure 2, under differentoperating modes

SYNC/MODE

LTC3411

PVIN

SWSVIN

PGOOD

PGOOD

ITHSHDN/RT

PGND

SGND

VFB

2.2µHVOUT2.5V/1.25A

VIN2.5V TO 5.5V

887k

100k

412k1000pF

3411 F01

22µF

13k

22µF

324k

C1, C2: TAIYO YUDEN JMK325BJ226MML1: TOKO A914BYW-2R2M (D52LC SERIES)

BURST MODEPULSE SKIPPING MODE

Figure 2. Step-down 2.5V/1.25A regulator


DESIGN FEATURES

tive choice for use in switching regu-lators. Unfortunately, the ESR is solow that it can cause loop stabilityproblems. Solid tantalum capacitorESR generates a loop zero at 5KHz to50KHz that is instrumental in givingacceptable loop phase margin. Ce-ramic capacitors remain capacitive tobeyond 300KHz and usually resonatewith their ESL before ESR becomeseffective. Also, ceramic caps are proneto temperature effects, requiring thedesigner to check loop stability overthe operating temperature range. Forthese reasons, great care must betaken when using only ceramic inputand output capacitors. The LTC3411helps solve loop stability problemswith its OPTI-LOOP phase compen-sation adjustment, allowing the useof ceramic capacitors. For details,and a process for optimizing compen-

sation components, see Linear Tech-nology Application Note 76.

A typical application for theLTC3411 is a 2.5V step-down con-verter using only ceramic capacitors,as shown in Figure 2. This circuitprovides a regulated 2.5V output, atup to 1.25A, from a 2.5V to 5.5Vinput. Efficiency for the circuit is ashigh as 95% for a 3.3V input asshown in Figure 3.

Although the LTC3411 is capableof operating at 4MHz, the frequencyin this application is set for 1MHz byR4 to improve the efficiency. Also, theavailability of capacitors and induc-tors capable of 4MHz operation islimited.

Figures 3 through 6 show the trade-off between noise and efficiency forthe different modes for the circuit.Figure 3 shows the efficiencies, whileFigures 4, 5 and 6 show the output

voltage and inductor current for dif-ferent operating modes.

Burst Mode operation is the mostefficient for low current loads, but it isalso generates the most complicatednoise patterns. Figure 4 shows howBurst Mode operation produces asingle pulse or a group of pulses thatare repeated periodically. By runningcycles in periodic bursts, the switch-ing losses—dominated by the gatecharge losses of the power MOSFET—are minimized. Figure 5 shows how inpulse skipping mode, the LTC3411continues to switch at a constantfrequency down to very low currents,minimizing the ripple voltage andripple current. Finally, Figure 6 showshow in forced continuous mode, theinductor current is continuouslycycled, creating a constant ripple atall output currents. Forced continu-ous mode is particularly useful innoise-sensitive telecom applicationssince the constant frequency noise iseasy to filter. Another advantage ofthis mode is that the regulator iscapable of both sourcing and sinkingcurrent into a load. This mode isenabled by forcing the mode pin tohalf of VIN.

Single Cell Li-Ion to 3.3VDC/DC ConverterLithium-Ion batteries are popular inmany portable applications becauseof their light weight and high energydensity, but the battery voltage rangesfrom a fully charged 4.2V down to adrained 2.5V. When a device requiresa voltage output that falls somewhere

Figure 4. Burst Mode operation

VOUT10mV/

DIV

IL1100mA/

DIV

VIN = 3.3VVOUT = 2.5VILOAD = 50mACIRCUIT OF FIGURE 2

2µs/DIV

Figure 5. Pulse skipping mode

VOUT10mV/

DIV

IL1100mA/

DIV


2µs/DIV

Figure 6. Forced continuous mode

VOUT10mV/

DIV

IL1100mA/

DIV


2µs/DIV

PVIN

LTC3411

PGNDSWSVIN

SGNDPGOODPGOODVFB SYNC/MODEITH SHDN/RT

L13.3µH D1 VOUT

3.3V/400mA

VIN

BM

VIN2.5V

TO 5V

100kM1

R4324k

R313k

3411 TA02

C31000pFC7

10pF

C1, C2: TAIYO YUDEN JMK325BJ226MM (408) 573-4150C4: SANYO POSCAP 6TPA47M (619) 661-6835D1: ON SEMICONDUCTOR MBRM120L (602) 244-6600L1: TOKO A915AY-3R3M (D53LC SERIES) (847) 699-3430

M1: SILICONIX Si2302DS (800) 554-5565

R1280k

C447µF

+

C122µF

C222µF×2

R2887k

Figure 7a. Single inductor, positive, buck-boost converter


DESIGN FEATURES

PVIN

LTC3411

PGOOD PGOODSWSVIN

SYNC/MODEVFBITH

SHDN/RT

L11µH

VOUT1.8VAT 1.25A

VIN2.5V

TO 4.2V

SGND PGND

R5100k

C4 22pF

R4154k

R315k

R1698k

R2887k

3411 TA04

C3470pF

C747pF

C51µF

+

C133µF

+C61µF

C1, C2: AVX TPSB336K006R0600 (207) 282-5111C4, C5: TAIYO YUDEN LMK212BJ105MG (408) 573-4150

L1: COILCRAFT DO1606T-102 (847) 639-6400

C233µF

Figure 8a. Tiny 1.8V/1.25A step-down converter uses low profile components

the middle of the Li-Ion operatingrange, say 3.3V, a simple buck orboost converter does not work. Onesolution is a single inductor, positivebuck-boost converter, which allowsthe input voltage to vary above andbelow the output voltage.

In Figure 7, the LTC3411 is used ina single Inductor, positive buck-boostconfiguration to supply a constant

3.3V with 400–600mA of load cur-rent, depending on the battery voltage.This circuit is well suited to portableapplications because none of the com-ponents exceed 3mm in height.

The efficiency varies with the inputsupply, due to resistive losses at highcurrents and to switching losses atlow currents. The typical efficiencyacross both battery voltage and loadcurrent is about 78%.

It’s Only 2mm High: 2MHz,Li-Ion to 1.8V ConverterIn some applications, minimizing theheight of the circuit takes prime im-portance. One method of lowering theDC/DC converter height is to run theLTC3411 at the 2MHz switching fre-quency, which allows one to uselow-profile inductors and capacitors.Figure 8 shows a circuit built with lowprofile components to produce a 2mmtall (nominal), 1.8V step-down con-verter that occupies less than 0.3

LOAD CURRENT (mA)

EFFI

CIEN

CY (%

)

100

95

90

85

80

75

70

65

60

55

501 100 1000 10000

3411 TA05

10

VOUT = 1.8VfO = 2MHz

2.5V

3.6V

4.2V

Figure 8b. Efficiency for thecircuit in Figure 8a

LOAD CURRENT (mA)10

EFFI

CIEN

CY (%

)85

80

75

70

65

60

55100k 1000

3411 TA03

VIN = 4V

VIN = 3VVIN = 3.5V

VIN = 2.5V

fO = 1MHz

Figure 7b. Efficiency for thecircuit in Figure 7a

square inches. In the spirit of keepingthings as small as possible, this cir-cuit uses tantalum capacitors for theirrelatively small size when comparedto equivalent ceramic capacitors.

The downside to running at a higherfrequency is that efficiency suffers alittle due to higher switching losses.The efficiency for this particular cir-cuit peaks at 93% with VIN = 2.5V.

ConclusionThe LTC3411 is a monolithic, step-down regulator that switches at highfrequencies, lowering componentcosts and board real estate require-ments of DC/DC converters. Althoughthe LTC3411 is designed for basicbuck applications, its architecture isversatile enough to produce anefficient single inductor, positive buck-boost converter, due in part to itspower saving Burst Mode operationand the OPTI-LOOP compensationfeature.

the target output voltage is reached,DONE goes high while CHARGE isalso high. The output of A1 goes high,which forces CHARGE high regard-less of the PWM signal. The part isnow in the Refresh mode. Once therefresh period is over, the DONE pingoes low, allowing the PWM signal todrive the CHARGE pin once again.This function can be easily imple-mented in a microcontroller. Figure 9shows the input current for the cir-

cuit of Figure 1 as the duty cycle of thePWM signal is varied.

ConclusionThe LT3420 provides a highly effi-cient and integrated solution forcharging photoflash capacitors. Manyimportant features are incorporatedinto the device, including automaticrefresh, tightly controlled currentsand an integrated power switch, thusreducing external parts count. The

LT3420 comes in a small, low profile,MSOP-10 package, making for a com-plete solution that takes significantlyless PC board space than more tradi-tional methods. Perhaps mostimportantly, the LT3420 provides asimple solution to a complicated highvoltage problem, freeing camera de-signers to spend time on otherimportant matters, like increasing thepixel count or adding new camerafeatures.

LT3420, continued from page 8


DESIGN FEATURES

IntroductionThe LTC3701 is an efficient, low in-put voltage, dual DC/DC controllerthat fits into the tight spaces requiredby the latest portable electronics. Ituses 2-phase switching techniquesto reduce required input capacitance(saving space and cost) and increaseefficiency. The versatile LTC3701 ac-cepts a wide range of input voltages,from 2.5V to 9.8V, making it usefulfor single lithium-ion cell and manymulticell systems. It can provide out-put voltages as low as 0.8V and outputcurrents as high as 5A. The 100%duty cycle allows low dropout formaximum energy extraction from abattery, and the optional Burst Modeoperation enhances efficiency at lowload currents. It also includes otherpopular features, such as a PowerGood voltage monitor, a phase-lockedloop, and an internal soft start. Itssmall 16-lead narrow SSOP packageand relatively high operating fre-quency (300kHz–750kHz) allow theuse of small, surface mount compo-nents, making for a compact overallpower supply solution.

OperationFigure 1 shows the LTC3701 used ina step-down converter with an input

of from 2.5V to 9.8V and two outputsof 2.5V at 2A and 1.8V at 2A. Figure2 shows its efficiency versus loadcurrent. The LTC3701 uses a con-stant frequency, current modearchitecture with the two controllersoperating 180 degrees out of phase.

During normal operation, each exter-nal P-channel power MOSFET isturned on every cycle when the oscil-lator for that controller sets a latchand turned off when the current com-parator resets the latch. The peakinductor current at which the currentcomparator resets the latch is con-trolled by the voltage on the ITH/RUNpin, which is the output of the erroramplifier. The VFB pin receives theoutput voltage feedback signal, whichis compared to the internal 0.8V ref-

erence by the error amplifier. Whenthe load current increases, it causesa slight decrease in VFB relative tothe reference, which, in turn, causesthe ITH/RUN voltage to increase untilthe average inductor current matchesthe load current.

2-Phase OperationThe LTC3701 offers the benefits of 2-phase operation, which include lowerinput filtering requirements, reducedelectromagnetic interference (EMI)and increased efficiency.

In a single phase dual switchingregulator, both top-side P-channelMOSFETS are turned on at the sametime, causing current pulses of up totwice the amplitude of those from asingle regulator to be drawn from theinput capacitor. These large ampli-tude pulses increase the total RMScurrent flowing into the input capaci-tor, requiring the use of moreexpensive input capacitors, and in-creasing both EMI and losses in theinput capacitor and input power sup-ply.

With 2-phase operation, the twochannels of the LTC3701 are oper-ated 180 degrees out of phase. Thiseffectively interleaves the current

Dual DC/DC Controller Brings 2-PhaseBenefits to Low Input VoltageApplications

+

+

SENSE1–

VFB1

ITH/RUN1

SGND

ITH/RUN2

VFB2

PLLLPF

SENSE2–

SENSE1+

VIN

PGATE1

PGND

PGATE2

PGOOD

EXTCLK/MODE

SENSE2+

1

3

2

4

6

5

7

8

16

15

14

13

12

11

10

9

LTC3701

220pF

220pF

10k

10k

D1, D2: IR10BQ015 L1, L2: LQN6C-4R7 M1, M2: FDC638P

78.7k

80.6k

100k

169k

4.7µH

47µF 10µF

47µF

0.03Ω

0.03Ω

4.7µH

D1

M1 SW1

M2 SW2

D2

3701 F01a

VOUT12.5V2A

VIN2.5V TO 9.8V

VOUT21.8V2A

GND

Figure 1. 2-phase step-down converter with an input of 2.5V to 9.8V and two outputs: 2.5V at 2A and 1.8V at 2A

by Jason Leonard

The LTC3701 offers thebenefits of 2-phase

operation, which includelower input filtering

requirements, reducedelectromagnetic

interference (EMI) andincreased efficiency.


DESIGN FEATURES

pulses coming from the switches,greatly reducing the amount of timewhere they overlap and add together.The dead bands in the input currentwaveform are “filled up,” so to speak.The result is a significant reductionin the total RMS input current, whichin turn allows for the use of lessexpensive input capacitors, reducesshielding requirements for EMI, andimproves efficiency. Figure 3 showsthe input waveforms for the circuit inFigure 1. The RMS input current issignificantly reduced by the inter-leaving current pulses. Of course, theimprovement afforded by 2-phase op-eration is a function of the dualswitching regulator’s relative dutycycles, which are dependent on theinput voltage VIN. Figure 4 shows howthe RMS input current varies forsingle-phase and 2-phase operationfor 2.5V and 1.8V regulators over awide input voltage range.

Burst Mode OperationThe LTC3701 can be enabled to enterBurst Mode operation at low loadcurrents by connecting the EXTCLK/MODE pin to VIN. In this mode, theminimum peak current is set as ifVITH/RUN = 1V, even though thevoltage at the ITH/RUN pin is at alower value. If the inductor’s averagecurrent is greater than the load re-quirement, the voltage at the ITH/RUN pin will drop as VOUT risesslightly. When the ITH/RUN voltagegoes below 0.85V, a sleep signal isgenerated, turning off the externalMOSFET and much of the LTC3701’sinternal circuitry. The load current isthen supported by the output capaci-

tor. When the ITH/RUN voltage goesabove 0.925V, the sleep signal goeslow and normal operation resumes.For frequency sensitive applications,Burst Mode operation can be inhib-ited by connecting the EXTCLK/MODE pin to ground. In this case,constant frequency operation is main-tained at a lower load current with alower output voltage ripple. If the loadcurrent is low enough, cycle skippingoccurs to maintain regulation.

FrequencySelection/Synchronization(Phase-Locked Loop)The LTC3701 operates at a constantfrequency between 300kHz and

750kHz. The frequency can be se-lected by forcing a voltage at thePLLLPF pin. Grounding the PLLLPFpin selects 300kHz, while tying it toVIN or a voltage greater than 2V se-lects 750kHz. Floating the PLLLPFpin selects 550kHz operation.

The LTC3701 can also be synchro-nized to an external clock source(300kHz to 750kHz) using theLTC3701’s true phase-locked loop.The clock signal is applied to theEXTCLK/MODE pin and an RC filteris connected between the PLLLPF pinand ground. Burst Mode operation isdisabled when synchronized to anexternal clock.

Run/Soft StartEither controller can be shutdown bypulling its respective ITH/RUN pinbelow 0.35V, which turns off mostcircuits associated with that control-

INPUT VOLTAGE (V)2

0

INPU

T CA

PACI

TOR

RMS

CURR

ENT

0.2

0.6

0.8

1.0

2.0

1.4

4 6 73701 F04

0.4

1.6

1.8

1.2

3 5 8 9 10

SINGLE PHASEDUAL CONTROLER

2-PHASEDUAL CONTROLER

VOUT1 = 2.5V/2AVOUT2 = 1.8V/2A

Figure 4. RMS input currentcomparison

LOAD CURRENT (mA)

60EFFI

CIEN

CY (%

) 80

100

50

70

90

1 100 1000 100003701 F01b

4010

VIN = 3.3V

VIN = 8.4V

VOUT = 2.5V

VIN = 4.2V

VIN = 6V

Figure 2. Efficiency vs load current

+

SENSE1–

ITH/RUN1

VFB1

SGND

VFB2

ITH/RUN2

PLLLPF

SENSE2–

SENSE1+

VIN

PGATE1

PGND

PGATE2

PGOOD

EXTCLK/MODE

SENSE2+

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC3701

C4220pF

R510k

R678.7k

R7169k

R10.03Ω

L14.7µH

C147µF

C210µF

VIN2.5V TO 9.8V

VOUT2.5V4A

R20.03Ω

L24.7µH

D1

M1

M2

D2

3701 TA03

C1: SANYO 6TPA47M (619) 661-6835C2: TAIYO YUDEN LMK325BJ106K-T (408) 573-4150

D1, D2: INTERNATIONAL RECTIFIER IR10BQ015 (310) 322-3331L1, L2: MURATA LQN6C-4R7 (814) 237-1431

M1, M2: SILICONIX Si3443DV (800) 554-5565R1, R2: DALE 0.25W (605) 665-9301

Figure 5. 2.5V–9.8V to 2.5V/4A 2-phase step-down converter operating at 550kHz

Figure 3. These input waveforms forthe circuit in Figure 1 show how 2-phase operation reduces ripple.Reduced ripple translates directly toless expensive input capacitors,reduced shielding requirements forEMI, and improved efficiency.

SW15V/DIV

SW25V/DIV

INPUTCURRENT

1A/DIV


DESIGN FEATURES

ler and holds its external MOSFEToff. If both ITH/RUN pins are pulledlow, the LTC3701 is shutdown anddraws only 9µA.

The LTC3701 has separate inter-nal soft start functions that alloweach output to power up gently. Themaximum allowed inductor currentis stepped up from 0 to 120mV/RSENSEin four equal steps of 30mV/ RSENSE,with each step lasting 512 clock cycles(just under 1ms per step at 550kHz).

Power Good Output VoltageMonitorA window comparator monitors bothoutput voltages and the open-drainPGOOD output is pulled low whenthe divided down output voltages arenot within ±8% of the reference volt-age of 0.8V.

2-Phase 2.5V/2A and 1.8V/2A Step-Down RegulatorFigure 1 shows a typical applicationof the LTC3701. This circuit suppliesa 2A load at 2.5V and a 2A load at1.8V with an input supply from 2.5Vto 9.8V. Due to the reduced inputcurrent ripple associated with 2-phaseoperation, only a single 10µF ceramic

+

+

SENSE1–

VFB1

ITH/RUN1

SGND

ITH/RUN2

VFB2

PLLLPF

SENSE2–

SENSE1+

VIN

PGATE1

PGND

PGATE2

PGOOD

EXTCLK/MODE

SENSE2+

1

3

2

4

6

5

7

8

16

15

14

13

12

11

10

9

LTC3701

C6 470pF

C4 220pF

C3 10µF

R1047k

R510k

10k

10nF

R678.7k

R880.6k

R9100k

R7249k

R10.025Ω L1A

L1B

• •

C147µF

C222µF

C547µF

C1, C5: SANYO 6TPA47M (619) 661-6835C2: TAIYO YUDEN JMK325BJ226MM (408) 573-4150C3: TAIYO YUDEN JMK316BJ106ML

D1, D2: INTERNATIONAL RECTIFIER IR10BQ015 (310) 322-3331L1A, L1B: COILTRONICS CTX5-2 (561) 752-5000

L2: MURATA LQN6C-4R7 (814) 237-1431M1, M2: SILICONIX Si3443DV (800) 554-5565R1, R2: DALE 0.25W (605) 665-9301

R20.03Ω

L24.7µH

M1

M2

100k

550kHz

D2

D1

VIN

3701 TA06

VIN2.7V to 4.2V

VOUT13.3V1A

GND

VOUT21.8V2A

Figure 6. Single cell Li-Ion to 3.3V/1A and 1.8V/2A DC/DC converter

input capacitor is required. The 0.03Ωsense resistors ensure that both out-puts are capable of supplying 2A witha low input voltage. The circuit oper-ates at the internally set frequency of550KHz. 4.7µH inductors are chosenso that the inductor currents remaincontinuous during burst periods atlow load current.

2-Phase Single Output 2.5V/4A Step-Down RegulatorIn addition to dual output applica-tions, the LTC3701 can also be usedin a single output configuration totake advantage of the benefits of 2-phase operation, as shown in Figure5. This circuit provides a 2.5V outputwith up to 4A of load current. In thiscase, 2-phase operation reduces boththe input and output current ripple,in turn reducing the required inputand output capacitances.

Single Cell Li-Ion to 3.3V/1A(Zeta Converter) and 1.8V/2AIn addition to step-down applications,the LTC3701 can also be used in azeta converter configuration that willdo both step-down and step-up con-versions, as shown in Figure 6. Thiscircuit delivers 1A at 3.3V (zeta con-verter) and 2A at 1.8V (step-downconverter) from an input of 2.7V to4.2V (Li-Ion voltage range). The cir-cuit takes advantage of the LTC3701’strue phase-locked loop by synchro-nizing to an external clock source.

ConclusionThe LTC3701 brings the benefits of 2-phase operation to low-voltage dualpower supply systems. It offers flex-ibility, high efficiency, and many otherpopular features in a small 16-pinnarrow SSOP package.

forthe latest information

on LTC products, visit

www.linear.com


DESIGN FEATURES

IntroductionThe LTC4400-1 and LTC4401-1 pro-vide RF power controller solutions forthe latest cellular telephones. Theyfeature very small footprints, lowpower consumption and wide fre-quency ranges while minimizingadjacent channel interference by care-fully controlling RF power profiles.The LTC4400-1 and LTC4401-1 areboth available in a low profile 6-pinThinSOT package, and require fewexternal parts. For example, whenused with a directional coupler, onlytwo resistors and two capacitors arerequired (Figure 1a, Figure 1b). Bothdevices require minimal power to op-erate, typically 1mA when enabledand 10µA when in shutdown.

The LTC4400-1’s 450kHz loopbandwidth is optimized for applica-tions involving fast turn-on (<2µs)and medium gain (200-300dB/V) RFpower amplifiers. The LTC4401-1’s250kHz loop bandwidth is optimizedfor slow turn-on (>2µs) and/or highgain (300-400dB/V) RF power ampli-fiers. The RF frequency range for bothparts is 800MHz to 2.7GHz and thesupply voltage range is 2.7V to 6.0V.This wide frequency and voltage rangeallow these products to be used in avariety of RF power control applica-tions including GSM/GPRS, PCS andTDMA. The LTC4400-1 and LTC4401-1 include an auto zero system thatrequires periodic updates betweensingle or multiple consecutive bursts.Therefore these power controllers arenot suitable for continuous time ap-plications.

Figure 2 shows the block diagramof the LTC4400/4401. When the partis in shutdown all circuitry except thereference is turned off and VPCA isheld at ground. When the part isenabled, the auto zero system samplesboth internal and external offsets.

After 10µs the auto zero system isdisabled; the sampled offset voltagecorrection factor is held on two inter-nal capacitors. A differential holdscheme is used to convert hold ca-pacitor voltage droop (due to leakagecurrents) to a common mode voltagedroop. This common mode voltagedroop is rejected by the auto zeroamplifier, resulting in greatly in-creased auto zero hold time. The autozero system improves temperaturedependent characteristics by remov-ing temperature offset voltage driftsfrom internal and external sources.

The external power control ramp isapplied 12µs after SHDNB is assertedhigh by the baseband microproces-

sor. When the ramp is applied, theVPCA voltage begins to rise. The RFpower amplifier turns on when VPCAreaches the RF power amplifier’sthreshold voltage. VPCA actuallystarts from 450mV. This start voltagereduces the time required to turn onthe RF power amplifier and is lowerthan power amplifier threshold volt-ages used in mobile radio applications.The power control loop is open untilthe RF power amplifier turns on andstarts supplying an RF output signal.While the loop is open, the VPCA risetime is limited by the LTC4400/4401bandwidth and the magnitude of thePCTL signal. A portion of the RF out-put voltage is fed back to theLTC4400/4401 RF pin. This signal isthen peak detected by an internalSchottky diode and capacitor. Thedetected voltage is applied to the nega-tive input of the loop amplifier therebyclosing the power control loop. Oncethe loop has closed, the RF outputsignal follows the power ramp signalat PCTL.

RF Detector PerformanceThe LTC4400 and LTC4401 incorpo-rate two features to improve detectordynamic range. An auto zero systemeliminates both internal offsets and

ThinSOT RF Power Controllers SaveCritical Board Space and Power inPortable RF Products

by Ted Henderson and Shuley Nakamura

VCC

SHDN

PCTL

1

5

2

6

4

3

RF

VPCA

GND

LTC4401-1

33pF

68Ω

SHDN

DAC

1.8GHzINPUT

900MHzINPUT

50Ω4401 TA01

0.1µFLi-Ion

VPC900MHzOUTPUT

1.8GHzOUTPUT

PA MODULE

BANDSELECT

Figure 1b. Typical power control block diagram

Figure 1a. The DC401A RF demo board. Thecircled area encloses the LTC4401 powercontroller (U4) and its required externalcomponents (C11, C12, C13 and R2).


DESIGN FEATURES

external power control DAC offsets.Secondly, a compression circuit al-lows for higher feedback signals atlower RF power levels to extend thepower detector range. The fullyintegrated detector has a small tem-perature coefficient as shown inFigure 3.

Measuring RF PowerAmplifier Rise TimesTo determine which LTC RF powercontroller fits a particular application,the designer must first understandthe RF power amplifier turn-on char-

acteristics. Figure 4 shows a recom-mended test setup.

A pulse generator is used to drivethe RF amplifier power control pin,with its duty cycle set to minimizepower dissipation (i.e. 1/8 duty cycle).Terminate the RF power control pinwith a 50Ω resistor to match thepulse generator and avoid ringing.With a square wave pulse at variousamplitudes, determine the RF outputpower response. Measure at severaloutput power levels since the risetime may be power level dependent.

Use a high frequency digital scopeto measure the RF output voltage

shape. Figure 5 shows a typical RFoutput voltage response. This wave-form consists of two regions, delayand ramp. The ramp time is mea-sured from the start of the RF outputto 90% of the final amplitude. Gener-ally the LTC4400 is used for amplifierswith total delay and ramp times <2µs;the LTC4401 is used for amplifierswith total times >2µs. Other factorssuch as power amplifier gains, cou-pler and antenna switch losses, mayalso impact this selection. Very highgain power amplifiers may requirethe LTC4401 independent of the re-sponse times.

–

+RF DET

–

+GM

80mV

270kHzFILTER

–

+

30k

22k 51k

30k

250Ω

28pF

33pF

100Ω

4401-1 BD

12Ω

150k

60µA 60µA

RF1

VCC

Li-Ion

6

5VPCA

GND

PCTL

3

RF PARF IN

AZ

AUTOZEROTXENB

+–

+–

CONTROL

4

SHDN

10µsDELAY

VREF

50Ω

2

68Ω

VREF

GAINCOMPRESSION

CLAMP

CC BUFFER38k

30k

33.4k 6k

30k

–

+

–

+

TXENB

LTC4401-1

VBG

Figure 2. LTC4400/4401 block diagram


DESIGN FEATURES

Powerful and Easy to UseDevelopment Tools OptimizePA ControlVersion 2 of the LT ramp-shapingprogram (LTRSv2.VXE) is availablefrom Linear Technology. Figure 6shows the program window ofLTRSv2.VXE in ramp-shaping mode.This program lets users generate, re-shape, and load ramp profilewaveforms onto the DC314A digitaldemo board. The DC314A digital demo

board provides regulated power sup-plies, control logic and a 10-bit DACto generate the SHDNB signal and thepower control PCTL signal. Flashmemory and a serial port interfaceare also included for updating DACprofiles stored on the DC314A. Eightpower control profiles can be stored

in the flash memory. A rotary switch(SW1) can be used to select the de-sired power profile. The DC314Aprovides signals to the DC401A RFdemo board, which contains a GSM/DCS RF channel, LTC4401-1 Thin-SOT power controller, and HitachiPF08107B power amplifier (Figure1B). The RF test measurement setupis shown in Figure 7.

LTRSv2.VXE creates smooth rampwaveforms based on user inputs. Theuser controls all aspects of the rampparameters such as initial DAC off-set, step voltage and time, rise andfall times, and maximum voltageamplitude and time. LTRSv2.VXEuses a raised-cosine function to cre-ate smooth transitions between areasof varying amplitudes, such as be-tween the step and the maximumamplitude (Figure 8).

LTRSv2.VXE ramp profile param-eters are saved in tables as text files.Linear Technology distributesLTRSv2.VXE with HP VEE Runtime,

SMA CABLERF

CARRIER 50Ω INPUT

50Ω INPUTHP 8116A

RF PA

COAXIAL CABLE

TRIGGEROUT

TEKTRONIXTDS820 6GHz

OSCILLOSCOPE

50Ω

VPC

COAXIALCABLE

Figure 4. RF power amplifier rise time testsetup

RF INPUT POWER (dBm)

10

PCTL

REF

EREN

CED

DETE

CTOR

OUT

PUT

VOLT

AGE

(mV)

100

1000

10000

–24 –12 –8 –4 40 8 12 161

–20 –164401 G03

75°C 25°C–30°C


10

PCTL

REF

EREN

CED

DETE

CTOR

OUT

PUT

VOLT

AGE

(mV)

100

1000

10000

–22 –10 –6 –2 62 10 141

–18 –144401 G04

75°C 25°C–30°C

Figure 3c. Detector characteristicsat 2400MHz

Figure 3d. Detector characteristicsat 2700MHz


10

PCTL

REF

EREN

CED

DETE

CTOR

OUT

PUT

VOLT

AGE

(mV)

100

1000

10000

–28 –10 2 81

–22 –4 14–16

75°C 25°C–30°C


10

PCTL

REF

EREN

CED

DETE

CTOR

OUT

PUT

VOLT

AGE

(mV)

100

1000

10000

–26 –14 –8 –2 4 10 161

–204401 G02

75°C 25°C–30°C

Figure 3a. Detector characteristicsat 900MHz

Figure 3b. Detector characteristicsat 1800MHz

Figure 3. Typical detector characteristics

Figure 5. RF output voltage

200ns/DIV

TRIGGER200mV/

DIV

RFOUTPUT

4V/DIV

Figure 6. LTRSv2.vxe program window


DESIGN FEATURES

HP I/O Libraries, and ramp profiletable templates for various poweramplifiers. Each ramp profile wave-form table can be edited or overwrittenusing LTRSv2.VXE. These ramp pro-file table templates serve as anexcellent starting place for ramp-shaping. Figure 9 is an illustration ofa typical ramp profile waveform withthe ramp parameters labeled. Figure10 shows the program window wherethe user changes ramp profile wave-form parameters.

Ramp shapes vary depending onwhich power controller and poweramplifier are being used. For example,power amplifiers that exhibit “slow”turn-on/off times (2µs and greater)require larger step amplitude and timevalues and a higher DAC offset volt-age. Similarly, rise and fall times forslow power amplifiers are longer.

Ramp-shaping is an iterative pro-cess. Changes should be made oneparameter at a time since each affectsdifferent aspects of the output. Plac-ing oscilloscope probes on PCTL andVPCA greatly facilitates the ramp-shaping process.

The ramp waveform begins withthe DAC offset voltage. The offsetimproves ramp down characteristicsof the power amplifier. A 100mV off-set voltage is sufficient for theLTC4400-1 and fast power amplifi-ers, while a 200mV offset voltage issufficient for the LTC4401-1 and slowpower amplifiers. The offset time forthe ramp is typically 12µs duringwhich auto zeroing occurs. Figure 11shows the timing relationship betweenSHDNB, VPCA and PCTL.

amplitude to effect a correspondingchange in the RF output power. Oncethe output power is set, adjust theinitial step amplitude and time.

The initial step values are respon-sible for closing the voltage loop. VPCAmust quickly rise to the RF poweramplifier threshold voltage in order tomeet power versus time specifica-tions. If the initial step time oramplitude values are too low, thecontrol voltage waveform will resembleVPCA in Figure 12. The resolution ofthe DAC allows for amplitude changesas small as 2mV. The step voltage canbe changed in 1/2 microsecond mul-tiples. There are some tradeoffs totake into consideration when choos-

The first step in ramp-shaping isdetermining the correct output power.Increase or decrease the maximum

Figure 10. Ramp profile waveform parameter editing window

Figure 8. Rise and fall ramp shapes with raised-cosine function

Figure 9. Typical ramp profile waveform

INITIALOFFSET

STEPAMPLITUDE

0VAMPLITUDE

MAX LEVEL AMPLITUDE

MAX LEVEL TIME

STEPTIME

RISETIME

FALLTIME

ZEROTIME

12µs

PCRUNNING

LTRSv2.VXE

DC314A-ASERIAL CABLEDC401A

5V, 3A

AGILENT3631A

AGILENTHP8594E*

SPECTRUMANALYZER

AGILENTE4433B

RF SIGNALGENERATOR

SHDNB

SMA CABLE

3dB ATTENUATOR

SMA CONNECTOR

SMA CONNECTOR

20dBATTENUATOR

SMACABLE

COAXIAL CABLE

EXTTRIGGERINPUT

* HP 85722B ANDHP 85715B FORDCS AND GSMMEASUREMENTPERSONALITIES

Figure 7. Demo board evaluation setup for GSM/DCS measurements


DESIGN FEATURES

ing which parameters to change. Forinstance, if the step amplitude is toohigh, the RF output spectrum mayexhibit spurs. However, if the steptime is too long, as shown in Figure13, meeting required power versustime is compromised because the timeallotted for the burst portion is insuf-ficient. Figure 14 shows the idealshape for the control voltage. The riseportion of VPCA is smooth and has aconstant slope until the maximumamplitude is reached.

Once the step values are set, therise and fall times should be adjusted.If the rise time is too short for slowamplifiers, an overshoot will occurand will be visible in the power versustime measurement. Lengthening thefall time generally lowers the spurs±400 kHz from the center frequency.Rise and fall times vary from 8µs–14µs.

The last step is adjusting the widthof the maximum amplitude. This isnecessary to meet power versus timespecifications. Typically, the burstportion of the width is 588µs. Thetotal time of the maximum rampamplitude must be enough to passthe power versus time measurementand leave suitable time at 0 volts toturn the power amplifier off. Usually,1µs is required to turn off a poweramplifier.

After each parameter is changed, agraph of the waveform created ap-pears in the program window alongwith the option to load the ramp ontothe DC314A demo board.

Ramp-shaping is more challeng-ing with slower power amplifiersbecause more time is required on thestep, rise and fall. If there is notenough time to meet the power versustime mask and turn off the PA, then itis necessary to change the step am-plitude and time. A change of 4mV to6mV accounts for 1µs. Be careful to

not let the step amplitude become toohigh to avoid spurs in the RF output.

Figure 15 shows the control volt-age waveform for maximum outputpower at 1800MHz (DCS0). The wave-form has an initial start voltage of450mV. By starting the output con-trol voltage at 450mV, the timerequired to reach the power amplifierthreshold voltage is reduced. The startvoltage is generated by the LTC4400/4401 and not by the program.

Figure 16 is the correspondingoutput RF spectrum for the controlvoltage shown in Figure 15. The cen-ter frequency is 1710.2MHz and theinput power to the power amplifier is0dBm. Figure 17 shows the powerversus time measurement. The on-screen table, shown in Figure 6,represents the values entered to cre-ate the ramp waveform. The inputstep and ramp amplitudes include a200mV offset amplitude. Therefore,the actual step voltage is 36mV andthe ramp amplitude voltage is 1.24V.

Figure 12. PCTL and VPCA waveformswith low ramp step amplitude andstep time

PCTL500mV/

DIV

VPCA500mV/

DIV

5µs/DIV

Figure 13. PCTL and VPCA waveformswith high ramp step amplitude andstep time

PCTL500mV/

DIV

VPCA500mV/

DIV

5µs/DIV

10µs 28µs2µs

28µs543µs

T2T1 T3 T4 T5 T6

VSTART

SHDN

VPCA

PCTL

4400 TA02

T1: PART COMES OUT OF SHUTDOWN 12µs PRIOR TO BURSTT2: INTERNAL TIMER COMPLETES AUTOZERO CORRECTION, <10µsT3: BASEBAND CONTROLLER STARTS RF POWER RAMP UP AT 12µs AFTER

SHDN IS ASSERTED HIGHT4: BASEBAND CONTROLLER COMPLETES RAMP UPT5: BASEBAND CONTROLLER STARTS RF POWER RAMP DOWN AT END OF BURSTT6: PART RETURNS TO SHUTDOWN MODE BETWEEN BURSTS

Figure 11. LTC4400/4401 timing diagram

Figure 15. Correct VPCA response and450mV start voltage

VPCA200mV/

DIV

5µs/DIV

Figure 14. Optimized PCTL input andVPCA output

PCTL500mV/

DIV

VPCA500mV/

DIV

5µs/DIV


DESIGN FEATURES

Directional CouplerAlternativesThe DC401A board contains theLTC4401-1 power controller andHitachi PF08107B dual-band poweramplifier as well as a Murata dualband directional coupler and Muratadiplexer (Figure 18). The directionalcoupler has a coupling loss of14±1.5dB for the DCS frequenciesand 19±1dB for the GSM frequencies.While the directional coupler is a vi-able solution, there is a cheaper andsmaller solution that is comparablein performance (Figure 19).

This new scheme completely elimi-nates the directional coupler, 50Ωtermination resistor, and 68Ω shuntresistor. Instead, the RF signal is feddirectly to the diplexer from the poweramplifier. The RF signal is coupled

back to the LTC4401-1 via a capaci-tor and a series resistor. Thecomponent count is reduced by two.

The series capacitor should be inthe range of 0.3pF to 0.4pF and havea tolerance of ±0.05pF or less. Thetolerance is important because it di-rectly affects how much RF signal iscoupled back to the RF pin on theLTC4400/4401. ATC has ultra-lowESR, high Q microwave capacitorswith the tolerances desired. TheATC 600S0R3AW250XT and ATC600S0R4AW250XT are 0.3pF and0.4pF capacitors with 0.05pF toler-ance. These capacitors come in a small0603 package. The series resistor is49.9Ω with 2% tolerance as shown inFigure 19. There are several factors toconsider when using this technique,such as board layout and loading in

Figure 17. Power versus timemeasurement for DCS0

REFERENCE = 32.0dBmCENTER = 1.710200 GHz SPAN = 0HzRESOLUTION BANDWIDTH = 300kHzVIDEO BANDWIDTH = 300kHzSWEEP = 800µs

REFERENCE = 30.0dBmCENTER = 1.710200 GHz SPAN = 4.00MHzRESOLUTION BANDWIDTH = 30kHzVIDEO BANDWIDTH = 100kHzSWEEP = 2.00s

Figure 16. Output RF spectrumswitching transients for DCS0

LTC4401-1

PF08107B

49.9Ω

0.4pF

RF OUTDIPLEXER

ANTENNA

RFINVPC

VPC

Figure 19. Block diagram of directionalcoupler alternative

the main line. For example, parasiticeffects can significantly alter the feed-back network characteristics.

ConclusionLinear Technology has introduced twonew controllers to its RF power con-troller family. The LTC4400-1 andLTC4401-1 represent small, low powersolutions for RF power control. Theintegration of the RF detector, autozero system and compensated loopamplifier have produced a tempera-ture stable RF power control solution.External and internal voltage offsetchanges due to temperature or powersupply are cancelled whenever thepart cycles through shutdown. Theseproducts are available in a small, lowprofile ThinSOT package and operateover a frequency range of 800MHz to2700MHz. The demo boards discussedhere and ramp-shaping software areavailable upon request. Demo boardsfeaturing power amplifiers made byAnadigics, Conexant, Hitachi andRFMD are also available.

Figure 18. DC401A RF demo board schematic

33pF

68Ω

SHDN

4401 TA01

100pF0.1µF

VAPC

GSM INPUT

RFOUTPUTSMA

RAMP

BSEL

1µF×2 0.1µF 330pF

×2

1000pF

33pFDCS INPUT

15pFPIN_GSM

VBATT

32

6

5

1

1

2

8

7

6

PIN_DCS

VCTL

1

3

4

GSMIN

DCSIN

TERMINATION

GSMOUT

COUPLING

DCSOUT

4

5

9

POUT_GSM

POUT_DCS

GND

7

8

5

1

3

P2

P1GND GND

RF

VPCA

VCC4

3

51Ω

2, 6 2, 4, 6

MURATADIRECTIONAL COUPLER

LDC21897M19D-078

MURATADIPLEXER

LFDP20N0020A

VCTL>2V = GSM0V = DCS

VDD1 VDD2GND

PCTL

SHDN P3

LTC4401-1

HITACHIPF08107B

5


DESIGN IDEAS

IntroductionThe LTC3830 and the LTC3832 arepin-to-pin compatible upgrades to theLTC1430—a popular IC for low volt-age step-down applications due to itssimplicity and high efficiency. TheLTC3830 and the LTC3832 removethe LTC1430’s frequency foldback atstartup, thus eliminating inrush cur-rent and resulting output overshoot.Other improvements over theLTC1430 include tighter gm distribu-

tion of the error amplifier and tightercurrent limiting. The LTC3832 is iden-tical to the LTC3830, except that itincorporates a 0.6V reference for theoutput feedback, a larger gm and adefault frequency of 300kHz (instead

of the 200kHz for the LTC3830), mak-ing it good match for very low outputapplications. The higher frequency ofthe LTC3832 also allows the use ofsmaller inductors and capacitors,making for a smaller overall solution.

Versatile LTC3830 and LTC3832 DeliverHigh Efficiency for Step-Down, Step-Upand Inverting Power Conversions

Q1, Q2: SILICONIX Si7440DP (800) 554-5565CIN, COUT: SANYO POSCAP 6TPB330M (619) 661-6835

LIN: SUMIDA CDEP105-1R3-MC-S (847) 956-0667

5.6k

37.4k1%

12.7k1%RC

15k

CC3300pF

C168pF

Q1

MBR0520

B320A

10µF

10µFQ2

0.47µF

MBR0520VIN

3.3VCIN330µF

+5mΩ

LIN1.3µH

PVCC2

IMAX

IFB

G1

PGND

GND

G2

FB

PVCC1

VCC

SS

FREQSET

SHDN

COMP

SENSE+

SENSE–

LTC3830

0.1µF0.1µF

2.2µF

10µF10Ω

+ COUT330µF×2

VOUT5V5A

SHDN

NC

NC

NC

Figure 2a. Schematic diagram of 3.3V to 5V synchronous boost converter

+

G1

IMAX

IFB

G2

PGND

GND

SENSE+

FB

VCC

SS

FREQSET

SHDN

COMP

PVCC212k

1k

0.1µF

10µF

470pF

LO1.3µH13A

Q2

NC

D1B320A

VOUT2.5V12A

VIN3.3V–8V

RA12.4k1%

RB12.7k1%

Q1

0.1µF

LTC3830

10µF

+2.2µF

DZMMSZ5242B

PVCC1

SENSE–

NC

+

CIN220µF

+

C168pF

0.01µF

10Ω

1k

CC1500pF

130k

RC18.2k

D3MBR0520LT1

5V

COUT180µF

RUN

Q1, Q2: SILICONIX Si7440DP (800) 554-5565CIN: SANYO POSCAP 10TPB220M (619) 661-6835

COUT: PANASONIC EEFUD0D181R (714) 373-7334LO: SUMIDA CDEP105-1R3-MC-S (847) 956-0667

Figure 1. Schematic diagram of 2.5V/12A synchronous step-down power supply

DESIGN IDEASVersatile LTC3830 and LTC3832Deliver High Efficiency for Step-Down,Step-Up and Inverting PowerConversions ................................. 21Wei Chen and Charlie Zhao

How to Use the LTC6900 Low PowerSOT-23 Oscillator as a VCO ......... 23Nello Sevastopoulos

Save Space and Expense byExtracting Two Lowpass FiltersOut of a Single LTC1563 ............. 25Doug La Porte

Tiny and Efficient Boost ConverterGenerates 5V at 3A from 3.3V Bus................................................... 28

Dongyan Zhou

Small, Portable Altimeter Operatesfrom a Single Cell ....................... 29Todd Owen

Simple Isolated Telecom FlybackCircuit Provides Regulation WithoutOptocoupler ................................ 30John Shannon

Space Saving Dual Output ±5V HighCurrent Power Supply Requires OnlyOne 1.25MHz Switcher and OneMagnetic Component ................... 31Keith Szolusha

Efficient DC/DC Converter ProvidesTwo 15A Outputs from a 3.3VBackplane ................................... 32David Chen

Design Low Noise Differential CircuitsUsing the LT1567 Dual AmplifierBuilding Block ............................ 34Philip Karantzalis

by Wei Chen and Charlie Zhao


DESIGN IDEAS

This article shows several designsusing the LTC3830 for step down,step up and inverting applications.The LTC3832 can be used in place ofthe LTC3830 in any of these designs.All that is required are some minoradjustments to the feedback resistordivider and the compensation RC com-ponent values.

12A High Efficiency StepDown Power Supply Converts3.3V–8V Input to a 2.5VOutputLTC3830/3832 are voltage mode syn-chronous buck controllers with twopowerful MOSFET drivers for boththe main MOSFET and a synchro-nous MOSFET. The RDS(ON) of the mainMOSFET is used to establish the cur-rent limit, thus eliminating the senseresistor and its associated power loss.The current limit and switching fre-quency can be programmed easilythrough external resistors.

3.6k

100ΩMBR0520

1k

Q1

Q2

VIN3.3V

1µF

0.01µF

0.1µF

0.1µF

10µF

LO1.3µH

37.4k1%

12.7k1%

10µF1µF

+

+

RC15k

CC1.5nF

C168pF

DZ8.2V

CIN330µF

COUT330µF

VOUT–5V5A

Q1, Q2: SILICONIX Si7440DP (800) 554-5565CIN, COUT: SANYO POSCAP 6TPB330M (619) 661-6835

LO: SUMIDA CDEP105-1R3-MC-S (847) 956-0667

PVCC1

G1

IMAX

IFB

G2

FB

PGND

GND

PVCC2

VCC

SS

FREQSET

SHDN

COMP

SENSE+

SENSE–

LTC3830

SHDN

NC

NC

NC

13V

Figure 3. Schematic diagram of 3.3V to –5V inverting converter

Figure 1 shows the schematic dia-gram of a 12A step down design basedon LTC3830. The input is 3.3V to 8Vand the output is 2.5V. To obtaindifferent output voltages, vary theratio of RA/RB. With only two tinyPowerPak SO8 MOSFETs and 300kHzswitching frequency, this designachieves close to 90% efficiency with5V input and 2.5V output. The overallfootprint of this design is less than1"×1.2", with all of the componentsplaced on the same side of the board.For higher output currents, simplyparallel more MOSFETs and use aninductor with a higher current rating.

5A Step Up Power SupplyConverts 3.3V to 5VAlthough intended for synchronousbuck applications, LTC3830 andLTC3832 can also be used in othercircuit topologies. Figure 2a shows asynchronous boost design usingLTC3830 converting 3.3V to 5V. Com-

Figure 2b. How to use the DC resistance of the boost inductor to control current limiting

5.6k

Q1

Q2

LIN

IMAX

IFB

G1

G2

LTC3830 0.1µF

CF1µF

+COUT

VOUT

VIN

RF10k

pared to a conventional boost con-verter, this design uses a low RDS(ON)N-channel MOSFET to implement thesynchronous rectification, thereforeimproving efficiency by 5% to 10%.The maximum output current is 8Awith only two PowerPak SO8MOSFETs. A current sense resistor isused for more accurate current limit-ing than can be achieved by sensingRDS(ON) of the MOSFET. One may alsouse the DCR of the inductor to imple-ment the current limit function, asshown in Figure 2b. RF and CF filtersout the AC voltage components of theinductor voltage to obtain the DCvoltage drop on the DC resistance ofthe inductor. This scheme eliminatesthe sense resistor and its associatedpower loss, but the response toovercurrent conditions is slower thana topology that uses a sense resistor.The delay time is determined by theproduct of RF • CF.

5A Inverter Converts3.3V to –5VThe LTC3830 and LTC 3832 can alsobe used in inverting applications.Figure 3 shows a synchronous buck-boost power supply which converts3.3V into –5V. The total VCC supplyvoltage in this design is the sum of theabsolute values of input and outputvoltages, which is about 8.3V; andthe PVCC1 voltage is the VCC voltageplus 5V, which is 13.3V. Since thesevoltage stresses are very close to themaximum voltage ratings for theLTC3830 and the LTC3832 (VCC(MAX)= 9V and PVCC1(MAX) = 14V), Zenerdiodes should be placed on VCC andPVCC1 pins to provide overvoltage pro-tection.

ConclusionThe LTC3830 and LTC3832 are ver-satile voltage mode controllers thatcan be used in variety of applicationsincluding step up, step down andvoltage inversion. Their integratedhigh current MOSFET drivers andprogrammable frequencies allow us-ers to minimize power loss and totalsolution size.


DESIGN IDEAS

IntroductionThe LTC6900 is a precision low poweroscillator that is extremely easy touse and occupies very little PC boardspace. It is a lower power version ofthe LTC1799, which was featured inthe February 2001 issue of this maga-zine.

The output frequency, fOSC, of theLTC6900 can range from 1kHz to20MHz—programmed via an exter-nal resistor, RSET, and a 3-statefrequency divider pin, as shown inFigure 1.

R kMHz

N fSETOSC

=

20

10 100101

••

, N = (1)

A proprietary feedback loop linear-izes the relationship between RSETand the output frequency so the fre-quency accuracy is already includedin the expression above. Unlike otherdiscrete RC oscillators, the LTC6900does not need correction tables toadjust the formula for determiningthe output frequency.

Figure 2 shows a simplified blockdiagram of the LTC6900. The LTC6900master oscillator is controlled by theratio of the voltage between V+ andthe SET pin and the current, IRES,entering the SET pin. As long as IRESis precisely the current through resis-

tor RSET, the ratio of (V+ – VSET) / IRESequals RSET and the frequency of theLTC6900 depends solely on the valueof RSET. This technique ensures accu-racy, typically ±0.5% at ambienttemperature.

As shown in Figure 2, the voltage ofthe SET pin is controlled by an inter-nal bias, and by the gate to sourcevoltage of a PMOS transistor. Thevoltage of the SET pin (VSET) is typi-cally 1.1V below V+.

Programming the OutputFrequencyThe output frequency of the LTC6900can be programmed by altering thevalue of RSET as shown in Figure 1and the accuracy of the oscillator willnot be affected. The frequency canalso be programmed by steering cur-rent in or out of the SET pin, asconceptually shown in Figure 3. Thistechnique can degrade accuracy as

the ratio of (V+ – VSET) / IRES is nolonger uniquely dependent on thevalue of RSET, as shown in Figure 2.This loss of accuracy will becomenoticeable when the magnitude ofIPROG is comparable to IRES. The fre-quency variation of the LTC6900 isstill monotonic.

Figure 4 shows how to implementthe concept shown in Figure 3 byconnecting a second resistor, RIN,between the SET pin and a groundreferenced voltage source VIN.

For a given power supply voltage inFigure 4, the output frequency of theLTC6900 is a function of VIN, RIN,RSET, and (V+ – VSET) = VRES:

fMHzN

kR R

V V

V

OSCIN SET

IN

RES

=

+−( )

+

+

10 20

11

•

•

• RINRSET

1

(2)

When VIN = V+ the output frequencyof the LTC6900 assumes the highestvalue and it is set by the parallelcombination of RIN and RSET. Alsonote, the output frequency, fOSC, isindependent of the value of VRES = (V+

– VSET) so, the accuracy of fOSC iswithin the datasheet limits.

How to Use the LTC6900 Low PowerSOT-23 Oscillator as a VCO

by Nello Sevastopoulos

V+1

2

3

51kHz ≤ fOSC ≤ 20MHz5V

5V

10k ≤ RSET ≤ 2M

0.1µF

6900 TA01

4

GNDLTC6900

SET

OUT

DIV OPEN÷10

÷100

÷1

Figure 1. Basic connection diagram

–

+

+–

1

3

GAIN = 1

V+

VBIAS

IRES

IRES

RSET

SET

GND

MASTER OSCILLATOR

PROGRAMMABLEDIVIDER (N)

(÷1, 10 OR 100)

VRES = (V+ – VSET) = 1.1V TYPICAL

IRES(V+ – VSET)ƒMO = 10MHz • 20kΩ •

THREE-STATEINPUT DETECT

GND

V+

2µA

6900 BD

2µA

OUT

DIVIDERSELECT

5

DIV42

+–

+–

Figure 2. Simplified block diagram


DESIGN IDEAS

When VIN is less than V+, and espe-cially when VIN approaches the groundpotential, the oscillator frequency,fOSC, assumes its lowest value and itsaccuracy is affected by the change ofVRES = (V+ – VSET). At 25°C VRES variesby ±8%, assuming the variation of V+

is ±5%. The temperature coefficient ofVRES is 0.02%/°C.

By manipulating the algebraic re-lation for fOSC above, a simplealgorithm can be derived to set thevalues of external resistors RSET andRIN, as shown in Figure 4:1. Choose the desired value of the

maximum oscillator frequency,fOSC(MAX), occurring at maximuminput voltage VIN(MAX) ≤ V+.

2. Set the desired value of theminimum oscillator frequency,fOSC(MIN), occurring at minimuminput voltage VIN(MIN) ≥ 0.

3. Choose VRES = 1.1 and calculatethe ratio of RIN/RSET from thefollowing:

RR

V Vff

V V

Vf

f

IN

SET

IN MAXOSC MAX

OSC MININ MIN

RESOSC MAX

OSC MIN

=

−( ) −

−( )

( )−

−

+ +( )

( )

( )( )

(

( )1

1

(3)

Once RIN/RSET is known, calculateRSET from:

RMHzN

kf

V V VR

R

VR

R

SETOSC MAX

IN MAX RESIN

SET

RESIN

SET

=

−( ) + +

+

10 20

1

• •( )

( )

(4)

Example 1: In this example, theoscillator output frequency has smallexcursions. This is useful where thefrequency of a system should be tunedaround some nominal value.

Let V+ = 3V, fOSC(MAX) = 2MHz forVIN(MAX) = 3V and fOSC(MIN) = 1.5MHz for

VIN=0V. Solve for RIN/RSET by equa-tion (3), yielding RIN/RSET = 9.9/1.RSET = 110.1kΩ by equation (4). RIN =9.9RSET = 1.089MΩ. For standard re-sistor values, use RSET = 110kΩ (1%)and RIN = 1.1MΩ (1%). Figure 5 showsthe measured fOSC vs VIN. The 1.5MHzto 2MHz frequency excursion is quitelimited, so the curve fOSC vs VIN islinear.

Example 2: Vary the oscillator fre-quency by one octave per volt. AssumefOSC(MIN) = 1MHz and fOSC(MAX) = 2MHz,when the input voltage varies by 1V.The minimum input voltage is halfsupply, that is VIN(MIN) = 1.5V, VIN(MAX)= 2.5V and V+ = 3V.

Equation (3) yields RIN/RSET = 1.273and equation (4) yields RSET =142.8kΩ. RIN = 1.273RSET = 181.8kΩ.

For standard resistor values, use RSET= 143kΩ (1%) and RIN = 182kΩ (1%).

Figure 6 shows the measured fOSCvs VIN. For VIN higher than 1.5V theVCO is quite linear; nonlinearitiesoccur when VIN becomes smaller than1V, although the VCO remains mono-tonic.

The VCO modulation bandwidth is25kHz that is, the LTC6900 will re-spond to changes in the frequencyprogramming voltage, VIN, rangingfrom DC to 25kHz.

Note:

All of the calculations above assume VRES = 1.1V,although VRES ≈ 1.1V. For completeness, Table 1shows the variation of VRES against various parallelcombinations of RIN and RSET (VIN = V+). Calculatefirst with VRES ≈ 1.1V, then use Table 1 to get abetter approximation of VRES, then recalculate theresistor values using the new value for VRES.

VIN (V)0 0.5 1 1.5 2 2.5 3

f OSC

(MHz

)

6900 F09

2.00

1.95

1.90

1.85

1.80

1.75

1.70

1.65

1.60

1.55

1.50

RIN = 1.1MRSET = 110kV+ = 3VN = 1

Figure 5. Output frequency vs inputvoltage

VIN (V)0 0.5 1 1.5 2 2.5 3

f OSC

(kHz

)

6900 F10

3000

2500

2000

1500

1000

500

0

RIN = 182kRSET = 143kV+ = 3VN = 1

Figure 6. Output frequency vs inputvoltage

RIN RSET VRES, V+ = 3V (VIN V+= ) VRES, V+ = 5V

20k 0.98V 1.03V

40k 1.03V 1.08V

80k 1.07V 1.12V

160k 1.1V 1.15V

320k 1.12V 1.17V VRES = Voltage across RSET

Table 1: Variation of VRES for various values of RIN RSET

V+1

2

3

5V+

5V0.1µF

6900 TA01

4

GNDLTC6900

SET

OUT

DIV OPEN÷10

÷100

÷1IPR

RSET

V+1

2

3

5fOSCV+

5VRSET

RIN

VRES

0.1µF

6900 TA01

4

GNDLTC6900

SET

OUT

DIV OPEN÷10

÷100

÷1VIN +–

+

–

Figure 3. Concept for programming via current steering Figure 4. Implementation of the concept shown in Figure 3


DESIGN IDEAS

IntroductionLowpass filters are required in sys-tems for a variety of reasons: to limitthe noise bandwidth, smooth out tran-sition edges or remove unwantedsignals. To make it easy for designersto use lowpass filters, Linear Tech-nology Corporation developed theLTC1563-2 and LTC1563-3, for whicha simple formula and a single resistorvalue set the cutoff frequency. TheLTC1563 features two 2nd orderbuilding block sections, which can becascaded to form a 4th order filter.Some applications, though, do notrequire the higher order filtering, butthey do require more filters. For theseapplications, the LTC1563 buildingblock sections can be used separatelyto produce a dual 2nd or 3rd orderfilter, thus saving the space and ex-pense of additional ICs.

The FilterCAD™ filter designprogram from Linear Technology Cor-poration also helps designers createcustom lowpass filters using LinearTechnology Corporation products.FilterCAD does not directly supportdual filters, but it can be tricked into

putting one together. This articleshows how to use FilterCAD and theLTC1563 to create a single IC duallowpass filter.

About the LTC1563The LTC1563 is designed to be aneasy-to-use 4th order lowpass filter.The LTC1563-2 provides a Butter-worth transfer function while theLTC1563-3 provides a Bessel responsewhen applied with six equally valuedresistors. The LTC1563 family is notlimited to these transfer functionsthough. One can generate nearly anyarbitrary fourth order transfer func-tion with the LTC1563 by using variedresistor values. For custom filtering,use FilterCAD to analyze the frequencyresponse and step response. Other-wise, using equally valued resistors,setting the cutoff frequency is simplya matter of choosing the appropriateresistor value:

R = 10k • (256kHz/fC)where fC = Cutoff Frequency

Figure 1 shows the LTC1563 cir-cuit topology. As mentioned above,the 4th order filter is obtained bycascading two 2nd order section build-ing blocks. The sections are similar,but not identical—their capacitor val-ues are different. Figure 1 shows theLTC1563 with the two sectionsconnected separately, instead of cas-caded, to form two 2nd order filters,or with the addition of two capacitors(one for each filter), two 3rd orderfilters. The rest of this article showshow to design this and similar duallowpass filters with the LTC1563.

Using FilterCAD to Design aDual Filter with the LTC1563The following procedure shows howto design a dual lowpass filter usingFilterCAD. The accompanying illus-trations show the design of a dual 3rdorder filter: one filter is a 3rd orderButterworth with a cutoff frequencyof 50kHz, and the other is a 3rd orderBessel with a cutoff of 100kHz. Thevalues can be modified to fit otherapplications.

Save Space and Expense byExtracting Two Lowpass FiltersOut of a Single LTC1563 by Doug La Porte

16

SHUTDOWNSWITCH

SHUTDOWNSWITCH EN

LP

20k

20k

AGND

SALPAINVA

R31

R21

R11

C2A

V–

V+

AGND

VIN1

VIN2

7

8 1

9

2

C1A

46

–

+

SBLPB

LTC1563-X

INVB

R32

R22

R12

C2B

AGNDAGND

VOUT2

VOUT1

11

C1B

1315

–

+

Figure 1. This block diagram shows how the LTC1563’s two filter sections can be hooked up separately to yield a dual filter from a single-IC.


DESIGN IDEAS

The first order of business is toidentify the filter order and transferfunction. This is determined by theusual parameters of passband band-width, attenuation requirement andstep response, though transfer func-tion selection is a classical engineeringtrade-off problem. The “ideal brickwall” filter has outstanding attenua-tion just beyond the passband butsuffers from a step response withlarge overshoot, substantial ringingand a long settling time. At the otherend of the spectrum, filters with idealstep responses tend to have poor at-tenuation just beyond the passband.Choosing the best transfer functionfor any specific application ultimatelyrequires a compromise. FilterCAD canhelp you decide, but you will need thevalues in Table 1 and a little trial anderror.

Table 1 lists the coefficients formost of the popular 2nd and 3rd

order lowpass filters. In the table, findthe coefficients for the filters that bestmatch your application needs. Then,enter the coefficients into FilterCADto see the frequency and step re-sponses of the filters. Here’s how:1. Launch FilterCAD.2. Select the Enhanced Design

option.3. Click Next. The Enhanced

Design window appears (Fig-ure 2).

4. In the Enhanced Design Win-dow, click Custom (for theResponse item).

5. Enter 0 for the Gain Frequency(Fg), indicating a lowpass filter.

6. Enter the filter coefficients fromTable 1 into the Coefficientstable in FilterCAD.

7. Enter the cutoff frequency in theCustom Fc box. Note that the fO

entered in step 6 is now multi-plied by the Custom Fc value.

FilterCAD can only evaluate onefilter at a time, so you will need toenter the coefficients for one filter,evaluate it, and then replace thosecoefficients for the other filter to evalu-ate it. If you put in the coefficients forboth filters, FilterCAD will assumeyou want the results of a compositefilter, which is not what we are inter-ested in here.

If you are designing a dual 2ndorder filter, you only need to enter onerow of coefficients for each filter. Fora 3rd order, you need two rows: one1st order (corresponding to anexternal RC) and one 2nd order (cor-responding to a 2nd order section ofthe LTC1563). That’s why in Table 1there are two rows of coefficients forthe 3rd order filters and only one rowfor the 2nd order filters.

For each 3rd order filter in thisexample, enter the first row of coeffi-cients, and choose LP1 (1st order) asthe coefficient type, corresponding tothe 1st order external RC lowpass.

Enter the second row of coeffi-cients, and choose LP (2nd order) asthe coefficient type, corresponding tothe 2nd order filter built into LTC1563.8. Evaluate the filter by clicking

the Frequency Response andStep Response buttons in theEnhanced Design window.

9. Adjust the coefficients to get theperformance you require.

10.Repeat for the second filter.

Figure 2. Enhanced Design window

Filter Type Bessel 12dB

Transitional Gaussian

6dB Transitional

Gaussian Butterworth 0.01dB Ripple

Chebyshev 0.1dB Ripple Chebyshev

0.5dB RippleChebyshev

Charactersitics

Passband Gain Flat, no ripple Flat, no ripple Flat, no ripple Flat, no ripple 0.01dB ripple 0.1dB ripple 0.5dB ripple

Attenuation Slope Poor, unselective ←→ Best, most selective Step Response Best, no overshoot ←→ Poor, most overshoot

Coefficients fO Q fO Q fO Q fO Q fO Q fO Q fO Q

2nd Order 1.2736 0.5773 – – – – 1.0000 0.7071 0.9774 0.7247 0.9368 0.7673 0.8860 0.8638

3rd Order 1.4530 0.6910 1.5352 0.8201 1.5549 0.8080 1.0000 1.0000 0.9642 1.1389 1.2999 1.3409 1.0689 1.7062

1.3270 – 0.9630 – 0.9776 – 1.0000 – 0.8467 – 0.9694 – 0.6289 –

Table 1: Coefficients for popular 2nd and 3rd order lowpass filters (fO normalized for a 1Hz cutoff frequency)


DESIGN IDEAS

VIN2

VOUT1

VOUT2

VIN1

11.Once you have determined thecoefficients of the two filters,enter them all in the EnhancedDesign window custom responsecoefficient table.

Figure 2 shows this window for the50kHz Butterworth and 100kHz Bes-sel example. The individual filters thatmake up the Butterworth and Besselfilters must be entered in a specificorder. That is, enter one filter with theLP1 section listed first. Then, enterthe second filter with its LP1 sectionfirst. At this point do not bother tolook at the frequency or step responseresults, unless you are somehow in-terested the combined 6th order filter.This is where some trickery comes in.FilterCAD does not directly supportdual filters, but it can still be used todesign the dual filter that we want.

The next step is to choose the partyou would like FilterCAD to use, inthis case the LTC1563-2.12.In the Enhanced Design window,

click the Implement button. TheEnhanced Implement windowappears (Figure 3) with thecoefficient table you entered inEnhanced Design.

13.Click the Active RC button.14.Select LTC1563-2 from the list

of available parts.Why not use the LTC1563-3? The

reason is that both the LTC1563-2and the LTC1563-3 have fO-Q limita-tions for the first building blocksection. The limitation is greater withthe LTC1563-3, so use the LT1563-2.

Check the order of the sections tomake sure that it hasn’t changed

from how you entered them in theEnhanced Design window. FilterCADusually leaves the order alone, butsometimes it shuffles things a little.To change the order of any offendingrows, click one of the rows to select it,and while pressing the Control key,click on another row to select it too.With both rows selected, click theSwap All button to swap the two rows.Figure 3 shows the Enhanced Imple-ment window.

The final step is to generate theschematic for the dual filter.15.Click the Schematic button in

the Enhanced Implement win-dow. The Schematic windowappears showing a single 6thorder lowpass filter. This is notexactly what we want, but it’seasy to fix.

16.Print the schematic.17.Fix the schematic.

Break out some Liquid Paper® anda pencil, and look at Figure 4. Theconnection from the first section tothe second section must be broken. Adab or two of Liquid Paper should dothe trick. Also, the inputs and out-puts must be labeled. In Figure 4,VIN1 and VOUT1 correspond to the50kHz Butterworth; VIN2 and VOUT2correspond to the 100kHz Bessel.

ConclusionFilterCAD does not directly supportsingle part dual filter design, but itcan still help you design a dual filterwith the LTC1563. The example inthis article illustrates that the proce-dure is a little tricky, but the endresult is a simple, compact and costeffective solution.

Figure 4. FilterCAD does not produce the exact schematic you want, but all you need are a fewsimple modifications to FilterCAD’s design.

Figure 3. Enhanced Implement window

Liquid Paper is a registered trademark of the GilletteCompany

forthe latest information

on LTC products, visit

www.linear.com


DESIGN IDEAS

SGND

ITH

RUN/SS

VFB

SYNC/MODE

SW

BG

PGND

TG

VOUT

LTC1700

1

2

4

3

5

10

8

9

6

7

+

L13.3µH

C322µF

C110µF

C4330µF6.3V

100pF

100pF

470pF

470pF 33k

52.3k1%

30.1k 1%

M2

M1

VIN2V TO 3V

VOUT3.3V/1A

DN280 F03

+ C268µF6.3V

C1: TAIYO YUDEN CERAMIC JMK316BJ106ML (408) 573-4150C2: AVX TAJB686K006R (207) 282-5111C3: TAIYO YUDEN CERAMIC JMK325BJ226MC4: SANYO POSCAP 6TPB330M (619) 661-6853L1: MURATA LQN6C (814) 237-1431

M1: SILICONIX Si9804 (800) 554-5565M2: SILICONIX Si9803

Figure 3. 2-cell to 3.3V, 1A boost regulator

IntroductionCircuits that require 5V remain popu-lar despite the fact that modernsystems commonly supply a 3.3Vpower bus, not 5V. The tiny LTC1700is optimized to deliver 5V from the3.3V bus at very high efficiency,though it can also efficiently boostother voltages. The small MSOP pack-age and 530kHz operation promotesmall surface mount circuits requir-ing minimal board space, perfect forthe latest portable devices. By takingadvantage of the synchronous recti-fier driver, the LTC1700 provides upto 95% efficiency. To keep light loadefficiency high in portable applica-tions, the LTC1700 draws only 180µAin sleep mode. The LTC1700 featuresa start-up voltage as low as 0.9V,adding to its versatility.

The LTC1700 uses a constant fre-quency, current mode PWM controlscheme. Its No RSENSE™ feature meansthe current is sensed at the mainMOSFET, eliminating the need for asense resistor. This saves cost, spaceand improves efficiency at heavy loads.For noise-sensitive applications,Burst Mode operation can be dis-abled when the SYNC/MODE pin ispulled low or driven by an externalclock. The LTC1700 can be synchro-nized to an external clock rangingfrom 400kHz to 750kHz.

3.3V Input, 5V/3A OutputBoost RegulatorFigure 1 shows a 3.3V input to 5Voutput boost regulator which cansupply up to 3A load current. Figure2 shows that the efficiency is greaterthan 90% for a load current range of

Tiny and Efficient Boost ConverterGenerates 5V at 3A from 3.3V Bus

by Dongyan Zhou

200mA to 3A and stays above 80% allthe way down to a 3mA load.

C2 is a tantalum capacitor provid-ing bulk capacitance to compensatefor possible long wire connections tothe input supply. In applicationswhere the regulator’s input is con-

SGND

ITH

RUN/SS

VFB

SYNC/MODE

SW

BG

PGND

TG

VOUT

LTC1700

1

2

4

3

5

10

8

9

6

7

+

L13.2µH

C322µF

C122µF

C4470µF

270pF

470pF

470pF 22k

R1316k0.1%

R2 100k 1%

M2

M1

VIN3.3V±10%

VOUT5V/3A

DN280 F01

+ C268µF6.3V

C1, C3: TAIYO YUDEN CERAMIC JMK325BJ226M (408) 573-4150C2: AVX TAJB686K006R (207) 282-5111C4: SANYO POSCAP 6TPB470M (619) 661-6853L1: SUMIDA CEP1233R2 (847) 956-0667

M1: INTERNATIONAL RECTIFIER IR7811W (310) 322-3331M2: SILICONIX Si9803 (800) 554-5565

Figure 1. 3.3V to 5V, 3A boost regulator

LOAD CURRENT (mA)

60EFFI

CIEN

CY (%

) 80

100

50

70

90

1 100 1000 10000DN280 F02

4010

VIN = 3.3VVOUT = 5V

Figure 2. Efficiency of the circuit inFigure 1



DESIGN IDEAS

Some sports enthusiasts want toknow altitude changes from an initialelevation. A small, lightweight, por-table altimeter is easy to design usingmodern micromachined pressuretransducers. Inverting barometricpressure and compensating fornonlinearities in air-pressure changeswith respect to altitude produces areasonably accurate altimeter.

Figure 1 shows a small, handheldaltimeter based on a micromachinedpressure transducer. The circuit takesadvantage of the inverse relationshipbetween air pressure and altitude.The aim of this circuit is to be small,lightweight, and portable. Accuracyis not paramount; errors as high as3%, such as a 300ft error at 10,000ftaltitude, are acceptable. The speed ofthe circuit is also not critical: Extremechanges in altitude in millisecondsmay prove fatal to whoever is at-tempting to read the output.

The heart of the altimeter is anNPC-1220-015-A-3L pressure trans-ducer. This 5k bridge provides 0mV to50mV of output voltage for a 0psi to15psi pressure range. To power thetransducer and signal-conditioningcircuitry, LT1307 (IC1), generates 5Vfrom a single AA battery, and a chargepump generates a –5V supply. Thepressure transducer is driven by IC3B(LT1490), which uses a reference volt-age and a setting resistor on thetransducer to generate appropriatedrive current.

The output of the transducer drivesan LT1167 instrumentation amplifier(IC2) which provides an initial gain of21. A nonlinear gain stage, composedof IC3A and associated components,then inverts the output of the instru-mentation to provide a voltage that isinversely proportional to air pressure.D4 and R1 introduce the nonlineargain, and the final output is directlyproportional to altitude.

R2 performs gain calibration in thesignal-conditioning circuitry. Thispotentiometer calibrates out any nor-mal variations in part tolerances andsets the altimeter for a 100mV changein output for every 1000ft of altitude.The circuit has some initial offset, aswell as an offset that is determined bybarometric pressure variations. Youcan use R3 to R5 to null this offset,giving a 0V to 1V output for 0ft to10,000ft of altitude.

Altimeter testing was performedusing a DeHavilland DHC-6 Twin Otterfor an ascent to 13,000ft, followed by

Small, Portable Altimeter Operatesfrom a Single Cell by Todd Owen

`

1

+

1

LG

5V

–5V

7

7

2

2

6

11

8

33 45

L

3

2

L

243k

6

5

D41N5711

56.2k 169k

10kR2

R1

R359k

R450k

R514.3k

0V TO 2V = 0 TO 20,000 FT

5V

8

1

4

4

1k

RSET

5

LUCAS NOVASENSORNPC-1220-015A-3L

TO4 1/2DIGITDVM

549k

2.43k

D1 TO D3: MOTOROLA MBR0520L (800) 441-2447L1: COILCRAFT D01608-103 (847) 639-6400

38.3k–5V

++

L110µH

–

+–

–+

+

–

+

IC2LT1167G = 21

IC3BLT1490

IC3ALT1490

L

4

6

3

1

100k

1000pF324k

1M

LT1004-1.2

36k

V125

5V

–5V

5

21.5VAA CELL

220µF

10µF

10µF 10µFD2

D3

D1

IN

SHDN

VCC

IC1LT1307

SW

FBGND

V125

V125

V125

Figure 1. To produce a reasonably accurate altimeter, conditioning circuitry invertsthe barometric pressure of a micromachined pressure transducer and compensates fornonlinearities in air-pressure changes with respect to altitude.

free descent—limited by the engineer'sparasitic drag—to 3000ft. Subsequentdeployment of an aerodynamic decel-erator (Precision Aerodynamics IcarusOmega 190) prevented engineer in-jury or circuit damage. Aircraft rentalfor testing is available at many localairports. Extensive instruction in freedescent and the use of aerodynamicdecelerators are highly recommendedbefore undertaking testing of thisnature. Contact USPA at (703) 836-3495 for further information.


DESIGN IDEAS

IntroductionThe LT1725 uses a proprietary tech-nique to regulate an isolated outputvoltage without an optocoupler, thusgreatly simplifying flyback converterdesign and reducing the componentcount. The result is reduced designtime, smaller space requirements,lower cost, and improved perfor-mance.

Traditional isolated flyback con-verters employ a secondary sidevoltage reference and error amplifierthat drive an optocoupler, which sendsthe control signals back to the pri-mary side. In addition to being partsintensive, this approach places anoptocoupler in the feedback loop,which introduces a host of designproblems. Optocouplers are poorlydefined components—their gain isvariable and subject to degradation

over time. They are also relativelyslow. Optocoupler shortcomings addconsiderably to the total converterdesign time and ultimately limit per-formance.

Consider instead the schematic ofFigure 1. This is a flyback converterbased on the LT1725. There are ex-tremely few components and yet ahigh level of functionality. This de-sign is short circuit proof and includesan input undervoltage lockout for in-creased reliability. The performanceof this converter is shown in Figure 2.Output voltage is regulated to within1% over a 2:1 input voltage rangewith 10% or greater load. No loadregulation is within 2% over a 2:1input voltage range. This is well withinthe typical requirement of 5% regula-tion.

Simple Isolated Telecom FlybackCircuit Provides Regulation WithoutOptocoupler

Circuit OperationThe LT1725 flyback controller is acurrent mode control IC. Currentmode operation provides for inherentline transient rejection and simpleloop compensation. Current modecontrollers have an “inner” fast cur-rent control loop and a slower “outer”voltage control loop. The inner currentloop has immediate pulse-by-pulsecontrol of the switching MOSFET M1.A normal switching cycle is as fol-lows. The MOSFET M1 is turned on tobegin the cycle. Once M1 is turnedon, the current in the primary wind-ing of the flyback transformer rampsup. When the primary current reachesa level determined by the value of thevoltage on the VC pin, M1 is turnedoff. The voltage on the VC pin is set bythe LT1725’s output voltage controlloop—the outer loop. Once M1 turns

by John Shannon

PGND

ISENSE

GATE

VC

FB

R290.2Ω

M1

T1

VCC

VOUT5V2A

UVLO3VOUT

SGNDRCMPCROCMPMENAB

LT1725

R3275k

R3347k

C1433pF

6 3 14

109

8

7

13 12 4 5

15

11 1

2

16

ENDLYSFSTOSCAP tON

C21nF50V

R93k

R1433Ω

R147k

R239Ω

R1039Ω

R13820k

R3047k

R551k

R53.01k

1%

R433.2k

C60.1µF

+

+

C8100µF

D712CWQ06FN

R1118Ω

C7470pFD1

BAS21

C10100pF

C315µF

R1230Ω

C9100pF C5

0.1µF

VIN36V TO 72V

–VIN

C130.47µF

C122µF

T1: COILTRONICS CTX02-14989 (561) 752-5000C8: TDK C5750X5R0J107M (408) 392-1400

C13: TDK C5750X7R2A155MM1: INTERNATIONAL RECTIFIER IRF620 (310) 322-3331

ISOLATION1500V

•

•

•

•

•

Figure 1. –48V to 5V 2A isolated flyback converter



DESIGN IDEAS

ADSL modems, disc drives, note-book computers, and other dataacquisition circuits require high cur-rent, ±5V power supplies withswitching frequencies greater than1.1MHz to avoid interfering with noisesensitive circuitry. Figure 1 shows avery simple, compact and efficientsolution that uses a single 1.25MHzLT1765EFE monolithic step-downswitching regulator and only onemagnetic component. This circuit canprovide ±5V supplies from a 12Vsource with greater than 1A capabili-ties on both rails. The LT1765EFE’sinternal 3A power switch saves spaceby eliminating the requirement for anexternal MOSFET and its traces. Typi-cal efficiency is 84%, as shown inFigure 2. An alternative option is touse two ICs, which means paying aheavy toll in board space, overall cost,and complexity.

The LT1765EFE uses current-mode control to regulate the positiveoutput with its step-down convertertopology. The off-the-shelf CTX5-1A(L1) transformer, which has a

greater than 3A current rating and a1:1 turns ratio, induces the samevoltage across the secondary windingas the primary winding and maintainsa –5V output. A high current-densityceramic coupling capacitor creates a

low-impedance path for current torun between the IC and the negativeoutput, maintaining excellent cross-regulation, as shown in Figure 3. The3A minimum switch-current limit ofthe LT1765EFE and the thermally

Space Saving Dual Output ±5V HighCurrent Power Supply Requires OnlyOne 1.25MHz Switcher and OneMagnetic Component

200 400 600 800 1000

4.9

5.0

5.1

5.2

5.3

5.4

5.5

5.6

1.5A POSITIVE OUTPUT CURRENT


250mA POSITIVE OUTPUT CURRENT

NEGA

TIVE

SUP

PLY

VOLT

AGE

(V)

NEGATIVE SUPPLY LOAD CURRENT (mA)12000

5.7

4.8

Figure 3. The negative (–5V) supplymaintains excellent regulation (±3%)over a wide range of output currentswithout the use of a negative supplyfeedback network.

EFFI

CIEN

CY (%

)

NEGATIVE SUPPLY LOAD CURRENT (mA)10k1

86

7710 100 1k

78

79

80

81

82

83

84

85



250mA POSITIVE OUTPUT CURRENT

Figure 2. The efficiency of the circuitin Figure 1 is typically greater than80%, and as high as 85% for varyingoutput currents.

MAX

IMUM

AVA

ILAB

LENE

GATI

VE S

UPPL

Y LO

AD C

URRE

NT (m

A)

POSITIVE SUPPLY LOAD CURRENT (mA)10k10

10k

10100 1k

100

1k

Figure 4. The available negative outputcurrent (±3% voltage regulation on –5Voutput) increases as positive supply(5V) current increases until switchcurrent or thermal limitation arereached.

OUTPUT5V1.5A

OUTPUT–5V1.1A

* L1 IS A SINGLE CORE WITH TWO WINDINGS CTX5-1AC6: 4.7µF, 6.3V, X5R, CERAMICD1, D3: DIODES INC. B220A (805) 446-4800

INPUT12V

GND

C80.22µF

C34700pF

C4100pF

D1

C54.7µF6.3VX5RCERAMIC

C6

C210µF25VX5R

CERAMIC

C14.7µF

16VX5R CERAMIC

D2CMDSH-3

D3

L1A*

L1B*

BOOST

LT1765

VIN VSW

FB

SHDN

GND VC

SYNC

R163.4k

R210.0k

R33.3k

Figure 1. This high-current dual output power supply conserves space by consolidating themagnetics into a single component (L1) and by using ceramic capacitors.

by Keith Szolusha



DESIGN IDEAS

IntroductionThe 3.3V DC bus has become popularfor broadband networking systems,where it is tapped for a variety oflower voltages to power DSPs, ASICsand FPGAs. These lower voltagesrange from 1V to 2.5V and often re-quire high load currents. To maintainhigh conversion efficiency, powerMOSFET conduction losses from thestep-down converters must be mini-mized. The problem is that the 3.3Vbus also brings with it frequent use ofsub-logic level MOSFETs. Such MOS-FETs have a relatively high RDS(ON),limiting the full-load efficiency of a

Efficient DC/DC Converter ProvidesTwo 15A Outputs from a 3.3VBackplane by David Chen

converter to around 85%. A moreefficient solution is to use logic-levelMOSFETs, which have very low RDS(ON)but require a 5V supply. The LTC1876allows the use of logic-level MOSFETsby combining a 1.2MHz boost regula-tor, which produces a 5V bias supplyfrom a 3.3V input, with two step-down controllers, which provide thelow voltage outputs. By integratingall three regulators in a single IC, theLTC1876 makes for efficient powersupplies that can be small and inex-pensive.

PGOOD

TG1

SW1

BOOST1

VIN

BG1

EXTVCC

INTVCC

PGND

BG2

BOOST2

SW2

TG2

RUN/SS2

AUXSD

AUXVIN

AUXPGND

AUXGND

RUN/SS1

SENSE1+

SENSE1–

VOSENSE1

FREQSET

STBYMD

FCB

ITH1

SGND

3.3VOUT

ITH2

VOSENSE2

SENSE2–

SENSE2+

AUXSGND

AUXVFB

AUXSW3

AUXSW3

36

35

34

33

32

31

30

29

28

27

26

25

24

23

22

21

20

19

1

2

3

4

5

6

7

8

9

10

11

12

13

14

15

16

17

18

LTC1876

D4CMDSH-3

L34.7µH

FSLB2520-4R7M 0.1µFC36 1µF 6.3V

1000pF

470pF

30.9k8.06k

12Ω

10.2k

C271µF6.3V

1k

8.25k

470pF470pF

VOSENSE2

OPTIONALREMOTE

SENSE

VOSENSE1

17.4k

12Ω

10Ω

5V

10k47k47k

470pF470pF6800pF

0.1µF0.1µF

6800pF 470pF

0.01µF0.01µF

20k

20k

10k

0.01µF 5.6k

1000pF

1µF6.3V

10Ω

12Ω

12Ω

C172.2µF10V

C1610µF10V

Q1Si4838

1µF6.3V

D2UPS840

Q2Si4838

Q3Si4838

Q4Si4838+ C21

10µF10V

C221µF6.3V

+

D1BAT54A

0.47µF

0.47µF

1µF6.3V

L20.6µH

CDEP134-0R6-H

D2UPS840

L10.6µH

CDEP134-0R6-H

0.002Ω

0.002Ω

+ 330µF6V×3

VIN3.3V

+ 220µF4V×3

2.5VAT15A

VOUT1

+ 330µF2.5V×3

1.8VAT15A

VOUT2

10Ω

Figure 1. An LTC1876 design converts 3.3V to 2.5V at 15A and 1.8V at 15A

IOUT1 = IOUT2 (A)0

OVE

RAL

L EF

FICI

ENCY

(%)

92

94

96

6 10

90

88

2 4 8 12 1514

86

84

Figure 2. High efficiency of thedesign in Figure 1


DESIGN IDEAS

Design ExampleFigure 1 shows a design that provides2.5V/15A and 1.8V/15A from a 3.3Vinput. Because the LTC1876 providesa 5V bias for MOSFET gate drive, avery low RDS(ON) MOSFET Si4838(2.4mΩ typical) can be used to achievehigh efficiency. Figure 2 shows thatthe overall efficiency is above 90%over a wide range of loads.

CURRENTTHROUGH Q1

5A/DIV

CURRENTTHROUGH Q3

5A/DIV

INPUT CURRENTFROM 3.3V SUPPLY

5A/DIV

Figure 3. Each switcher has 5A peak current, butthe total ripple at the input is still only 5A,minimizing CIN requirements.

1.25µs/DIV

Figure 2 also shows that the lightload efficiency of this design is morethan 84%. This is a direct benefit ofthe Burst Mode operation of theLTC1876. Further efficiency improve-ments come from operating the twostep-down channels out-of-phase. Thetop MOSFET of the first channel isfired 180° out of phase from that ofthe second channel, thus minimizingthe RMS current through the input

capacitors. This significantly reducesthe power loss associated with theESR of input capacitors. Figure 3shows detailed current waveforms ofthis operation.

ConclusionThe LTC1876 uses three techniquesto efficiently power low voltage DSPs,ASICs and FPGAs from a low inputvoltage. The first technique uses aninternal boost regulator to provide aseparate 5V for the MOSFET gatedrive. Secondly, its Burst Mode op-eration achieves high efficiency atlight loads. Lastly is the out-of-phasetechnique which minimizes input RMSlosses and reduces input noise. Com-plete regulator circuits are kept smalland inexpensive, because all threeswitchers (one step-up regulator andtwo step-down controllers) are inte-grated into a single IC. For systemswhere a separate 5V is available orthe input supply is greater than 5V,the internal boost regulator can beused to provide a third step-up out-put with up to 1A switch current.

OUTP

UT V

OLTA

GE

OUTPUT CURRENT (mA)25000

5.25

4.75500 1000 1500 2000

4.8

4.85

4.9

4.95

5

5.05

5.1

5.15

5.2

VIN = 36V

VIN = 48V

VIN = 72V

Figure 2. LT1725 regulation

EFFI

CIEN

CY (%

)

OUTPUT CURRENT (mA)25000 500 1000 1500 2000

10

20

30

40

50

60

70

80

90

100

0

VIN = 48V

VIN = 36V

VIN = 72V

Figure 3. Efficiency vs outputcurrent for the circuit in Figure 1

off, the current that had been flowingin the primary of the transformerbegins to flow in the secondary. Thevoltage on the drain of M1 rises to alevel determined by the transformerturns ratio and the output voltage.Similarly, the voltage on the feedbackwinding rises to a level set by theoutput voltage. The LT1725 reads thevoltage on the feedback winding dur-

ing the flyback pulse using a propri-etary sampling technique. Thissampled voltage is then compared aprecision internal reference and cur-rent is added to or subtracted fromthe capacitor on the VC pin. This hasthe effect of modifying the M1 turn-offcurrent in such a way as to regulatethe output voltage. An important ben-efit of this sampling technique is that

output voltage information arrives atthe controller about a microsecondafter the switching cycle is terminated.In a conventional optocoupler-baseddesign. Delays of tens to hundreds ofmicroseconds occur in the optocou-pler alone, severely limiting theconverters transient response. Addi-tionally the LT1725 features internalslope compensation. This suppressessub-harmonic oscillations that canoccur with less sophisticated currentmode controllers. Sub-harmonic os-cillations increase output voltageripple and increase switching stress.

ConclusionThe LT1725 isolated flyback control-ler greatly simplifies the design ofisolated flyback converters. Comparedto traditional opto-isolated designs,an LT1725 based circuit has far fewercomponents, superior transient re-sponse and is easier to stabilize.



DESIGN IDEAS

IntroductionMany communications systems usedifferential, low level (400mV – 1Vpeak-to-peak), analog baseband sig-nals, where the baseband circuitryoperates from with a single low volt-age power supply (5V to 3V). Anydifferential amplifier circuit used forbaseband signal conditioning musthave very low noise, and an outputvoltage swing that includes most ofthe power supply range for maximumsignal dynamic range. The LT1567, alow noise operational amplifier(1.4nV/√Hz voltage noise density) anda unity-gain inverter, is an excellentanalog building block (see Figure 1)for designing low noise differentialcircuits. The typical gain bandwidthof the LT1567 amplifier is 180MHzand op amp slew rate is sufficient forsignal frequencies up to 5MHz. TheLT1567 operates from 2.7V to 12Vtotal power supply. The output volt-age swing is guaranteed to be 4.4Vand 2.6V peak-to-peak, at 1k loadwith a single 5V and 3V power supplyrespectively. The LT1567 is availablein a tiny MS8 surface mount package.

A Single-Ended ToDifferential AmplifierFigure 2 shows a circuit for generat-ing a differential signal from asingle-ended input. The differentialoutput noise is a function of the noiseof the amplifiers, the noise of resis-tors R1 and R2 and the noisebandwidth. For example, if R1 and R2are each 200Ω, the differential volt-age noise density is 9.5nV/√Hz and ina 4MHz noise bandwidth the totaldifferential noise is 19µVRMS (with alow level 0.2VRMS differential signal,the signal-to-noise ratio is an excel-lent 80.4dB). The voltage on Pin 5(VREF) provides flexible DC bias for thecircuit and can be set by a voltage

divider or a reference voltage source(with a single 3V power supply, theVREF range is 0.9V ≤ VREF ≤ 1.9V). Ina single supply circuit, if the inputsignal is DC coupled, then an inputDC voltage (VINDC) is required to biasthe input within the circuit’s linearregion. If VINDC is within the VREFrange, then VREF can be equal to VINDCand the output DC common modevoltage (VOUTCM) at VO1 and VO2 isequal to VREF. To maximize theunclipped LT1567 output swing how-ever, the DC common mode output

voltage must be set at V+/2. In addi-tion, the input signal can be ACcoupled to the circuit’s input resistorR1 and VREF set to the DC commonmode voltage required by any follow-ing circuitry (for example the input ofan I and Q modulator).

A Differential Buffer/DriverFigure 3 shows an LT1567 connectedas a differential buffer. The differen-tial output voltage noise density is7.7nV/√Hz. The differential buffercircuit of Figure 3, translates theinput common mode DC voltage(VINCM) to an output common modeDC voltage (VOUTCM) set by the VREFvoltage (VOUTCM = 2 • VREF – VINCM). Forexample, in a single 5V power supplycircuit, if VINCM is 0.5V and VREF is1.5V then VOUTCM is 2.5V.

A Differential to Single-Ended AmplifierFigure 4 shows a circuit for con-verting a differential input to asingle-ended output. For a gain equal

Design Low Noise Differential CircuitsUsing the LT1567 Dual AmplifierBuilding Block by Philip Karantzalis

3

5

8 4

7

2–

+ –

+

150Ω

V+ V– LT1567DN194 F01DN194 F01

7pF

600Ω600Ω

61

Figure 1. LT1567 analog building block

3

5

8 4

7

2–

+ –

+

150Ω

R2R1VIN

V+VREF

V+

V– LT1567DN194 F02

7pF

0.1µF

0.1µF

600Ω600Ω

VO2

VO1

61

GAIN =

VO1 = – GAIN • VIN + (GAIN + 1) • VREFVO2 = –VO1 + 2 • VREFVDIFF = VO2 – VO1VDIFF = 2 • GAIN • (VIN – VREF)

= VO1VIN

R2R1

Figure 2. A single-ended input to differential output amplifier


DESIGN IDEAS

3

5

8 4

7

2–

+ –

+

150Ω

604Ω 604ΩV1

V2

V+VREF

V+

V– LT1567DN194 F03

7pF

0.1µF

0.1µF

600Ω600Ω

VO2

VO1

61

VO1 = –V1 + 2 • VREFVO2 = –V2 + 2 • VREFVDIFF = VO2 – VO1 = V1 – V2OUTPUT DC COMMON MODEVOLTAGE, VOCM = 2 • VREF – VINCM

Figure 3. A differential input and output buffer/driver

3

5

8 4

7

2–

+ –

+

150Ω

R1 R2

R3 = R1

V1

V2

V+VREF

V+

V– LT1567

DN194 F04

7pF

0.1µF

0.1µF

C

600Ω600Ω

VOUT

61

GAIN =

VO = GAIN (V2 – V1) + VREF

f–3dB BANDWIDTH AT VOUT =

IF R1 = R3 = 604Ω, THEN

NOISE AT VOUT = GAIN • Vη • √fηBW

Vη IS THE INPUT REFERRED DIFFERENTIAL VOLTAGE NOISEDENSITY IN nV/√Hz

fηBW = 1.57 • f–3dB

≤ 5MHz

, R3 = R1R2R1

R2604Ω1.21k2.43k

Vη9.08.48.1

GAIN124

12 • π • R2 • C

Figure 4. A differential input-to-single-ended output amplifier

to one (R1 = R2 = 604Ω and VOUT = V2– V1) the input referred differentialvoltage noise density is 9nV/√Hz anddifferential input signal-to-noise ra-tio is 80.9dB with 0.1VRMS input signalin a 4MHz noise bandwidth. The in-put AC common mode rejectiondepends on the matching of resistorsR1 and R3 and the LT1567 invertergain tolerance (common mode rejec-tion is at least 40dB up to 1MHz withone percent resistors and two percentinverter typical gain tolerance). If thedifferential input is DC coupled, thenVREF must be set equal to input com-mon mode voltage (VINCM) (if VREF isgreater than VinCM then a peak volt-

age on Pin 7 may exceed the outputvoltage swing limit). The DC voltage atthe amplifier’s output (VOUT, Pin 1) isVREF.

ConclusionWith one LT1567 and two or threeresistors, it is easy to design low

noise, differential circuits for signalsup to 5MHz. The LT1567 can also beused to make of low noise second andthird order lowpass filters and secondorder bandpass filters with differen-tial outputs. See www.linear.com fora spreadsheet-based design tool forjust this purpose.

Figure 4. Efficiency of the circuitin Figure 3

EFFI

CIEN

CY (%

)

LOAD CURRENT (mA)1k1

100

4010 100

50

60

80

70

90

VIN = 2V

VIN = 2.5VVIN = 3V

VOUT = 3.3Vnected very close to a low impedancesupply, this capacitor is not needed.

2-Cell Input, 3.3V/1A OutputRegulatorIn digital cameras and other battery-powered devices, the LTC1700 makesfor a high efficiency boost regulator ina small package. Figure 3 shows a 2-alkaline cell to 3.3V output circuit.This circuit can supply 1A maximumoutput current. Figure 4 shows theefficiency at different battery volt-ages. Efficiency of this circuit peaksat 93%. If a lower RDS(ON) MOSFET(such as Si6466) is used for M1, the

LTC1700, continued from page 28maximum output current can be in-creased to 1.4A with about a 2%reduction in efficiency due to theincrease in gate capacitance.MOSFETs with lower than 2.5V gatethreshold voltages are recommended.The LTC1700 is also an ideal devicefor single cell Li-Ion battery to 5Vapplications.

ConclusionThe LTC1700 boost controller bringshigh efficiency and small size to lowvoltage applications. Its features areideally suited to both battery-poweredand line-powered applications.


NEW DEVICE CAMEOS

LTC1706-85 Meets theRequirements of Intel VRM8.5 Specification for LaptopsLinear Technology announces therelease of the LTC1706-85 VID Volt-age Programmer to address therequirements of the new Intel® VoltageRegulator Module 8.5 specification.The LTC1706-85 is a precision, digi-tally-programmed resistive dividerwhich adjusts the output of any 0.8Vreferenced regulator. The LTC1706-85 adheres to Intel’s VRM 8.5specification and can provide volt-ages from 1.05V to 1.825V in 25mVincrements based on the state of theVID inputs. With an exceptionallytight output accuracy of ±0.25%, theLTC1706-85 in turn relaxes the ac-curacy requirements on the associatedDC/DC converter while still allowingthe VRM to easily meet the Intel out-put specification.

The LTC1706-85 has been designedto work over a wide range of inputvoltages. Each VID input contains aninternal 40kΩ pullup resistor with anintegrated blocking diode. The inputsassume a high state if left uncon-nected, and it is acceptable to drivethe VID inputs with as much as 5Vwhile powering the LTC1706-85 fromas little as 2.7V, increasing designflexibility.

The LTC1706-85 has a versatilearchitecture that allows it to be teamedwith a wide variety of Linear Technol-ogy DC/DC converters. This flexibilitymakes a large selection of regulatorsolutions available to power Intel andother microprocessors. For instance,the LTC1706-85 and the LTC1778can be used together to create a high-efficiency power supply that cantolerate the high input voltage re-quirement of laptop computers. Forextremely high current applications,such as servers and workstations,one can use a single LTC1706-85 toprogram up to six LTC1629 polyphasehigh efficiency step-down DC/DCconverters. The end result is an ex-

tremely compact, powerful, and pro-grammable power supply that reliesonly on surface-mount components.

The LTC1778 and LTC1629 aretwo of the many 0.8V reference DC/DC converters that are served by theLTC1706-85. It also works well withthe LTC1622, LTC1628, LTC1702,LTC1735, LTC1772, LTC1773,LTC1929 or LTC3728. In addition,Linear Technology Corp. offers theLTC1709-85 as a single-chip solutionto the Intel VRM 8.5 specification.

The LTC1706-85 comes in a min-iature 10-lead MSOP package and isspecified to operate from –40°C to85°C.

Versatile Op Amp FamilyBrings Low Noise and HighSpeed to Low VoltageApplicationsThe LT1722, LT1723, and LT1724are single, dual, and quad opera-tional amplifiers that offer a uniquecombination of high performance fea-tures and low supply current.Performance is fully specified for op-eration from +5V or ±5V supplies.Each amplifier typically draws a mere3.7mA, yet offers 200MHz gain-band-width-product and 70V/µs slew rate.The amplifiers are unity-gain stable,even while driving capacitive loads upto 100pF. The low input noise densi-ties of 3.8nV/√Hz and 1.2pA/√Hzallow designers to build sensitive,high-speed preamps previously re-quiring amplifiers with 12V–15V splitsupplies.

The DC characteristics are as im-pressive as the AC properties,including guaranteed sub-millivoltoffset and low residual bias-cancelledinput current under 350nA. The bias-cancellation also eliminates the needfor matched source resistances, re-ducing the resistor count in manydesigns.

The LT1722 single op amp is avail-able in the compact SOT-23 5-lead,or a SO style 8-lead surface-mountpackage. The LT1723 dual is avail-

able in the SO 8-lead package. TheLT1724 quad comes in the 14-leadSO package. Each device is availablein both commercial and industrialtemperature range versions.

With the versatility of having low-noise and offset, along with highgain-bandwidth and packaging op-tions, the LT1722, LT1723, andLT1724 can provide optimal solu-tions for many sensor preamp, linereceiver, line driver, and other appli-cations where high speed, precision,and signal fidelity are key require-ments.

Multiprotocol TransceiverFamily Works from Single3.3V SupplyThe LTC2844, LTC2845 and LTC2846are a new family of multiprotocoltransceivers designed to operate froma single 3.3V supply and interfacewith 3.3V logic. These devices are the3.3V counterparts to the 5V LTC1544,LTC1545 and LTC1546. When com-bined with the LTC2844 or LTC2845,the LTC2846 forms a complete soft-ware configurable DTE or DCEinterface that supports RS232,RS449, EIA530, EIA530-A, V.35, V.36and X.21 protocols. Unlike 3-chipsolutions from other manufacturers,cable termination resistors are pro-vided on-chip. The chip set supportsV.11 data rates of up to 10Mb/s andV.28 modes of 128kbps and the re-ceivers feature failsafe operation inall modes.

The desired protocol is selected viathree mode pins, M0, M1 and M2,which can be driven by a micropro-cessor, or they can be hardwired inthe connector (allowing the protocolto be selected by simply plugging inthe appropriate cable). The DCE/DTEpin allows the microprocessor to con-figure a port as a DCE or DTE port.This pin may also be hardwired to fixthe port as a DCE or DTE or to makethe selection when the appropriatecable is plugged in.

The LTC2846 consists of a boostswitching regulator, a charge pump,three configurable drivers, threeconfigurable receivers and precisionresistor termination networks. It

New Device Cameos

Intel is a registered trademark of Intel Corporation


NEW DEVICE CAMEOS

serves to handle data and clock sig-nals in the DTE or DCE interface andprovides the necessary terminationfor each protocol. It also providespower to the LTC2844 or LTC2845companion chip and generates the 5Vand ±8V voltage levels needed for thevarious protocols. The LTC2844 orLTC2845 handle the control signalsin the DTE or DCE interface. TheLTC2844 has four drivers and fourreceivers and provides for an optionallocal loop-back test signal. TheLTC2845 has five drivers and fivereceivers and allows users to addremote loop-back as well as test modesignals.

The LTC2844 is available in a 28-lead SSOP package, while theLTC2845 and LTC2846 are packagedin 36-lead SSOP packages. Both in-dustrial and commercial grades areoffered. The LTC2844/LTC2846 andLTC2845/LTC2846 chip sets are inthe process of being certified for NET1,NET2 and TBR2 compliance.

Accurate, Low Power andFast 80MHz AmplifiersProvide Best Solution forLow Voltage SignalConditioningThe LT1801 and LT1802 are dual andquad, low power, high speed rail-to-rail input and output operationalamplifiers. The LT1801 and LT1802amplifiers consume a mere 2mA maxsupply current per amp and still pro-vide 80MHz gain-bandwidth productand DC accuracy required by lowvoltage signal conditioning and dataacquisition systems.

The DC performance is exceptionalwith a maximum input offset voltageof 350µV and input bias current of250nA. These results come from in-ternal trimming of the input offsetvoltage and employing Linear Tech-nology Corporation’s patent pendingtechnique of the input bias currentcancellation.

The LT1801 and LT1802 have thecharacteristics that are essential inprecision systems: common moderejection of 105dB, power supply re-jection of 97dB and an open loop gainof 85V/mV combine to maintain pre-

cision performance over a wide com-mon mode input voltage, independentof power supply fluctuation, with mini-mum gain error.

These amplifiers can operate fromsupplies as low as 2.3V over indus-trial temperature ranges; have aninput voltage range that includes bothpower supply rails; and have an out-put that swings within 20mV of eithersupply rail to maximize the signaldynamic range in low voltage applica-tions. The rail-to-rail input and outputcharacteristics of the amplifiers cansimplify designs by eliminating a nega-tive supply. The LT1801 and LT1802also possess an 80MHz gain band-width product, a 25V/µs slew rateand a 50mA output current that makethem suitable for high frequency sig-nal conditioning. In servo loopapplications, where avoiding phasereversal is critical, the inputs of theseamplifiers can be driven beyond sup-plies without phase reversal of theoutput.

The LT1801 is housed in an SO-8package; the LT1802 in an SO-14—both with standard op amp pin outs.

Tiny TSOT-23 BuckRegulators Are Optimized toWork with Ceramic OutputCapacitors for Very LowOutput RippleThe LTC3405A, LTC3405A-1.5 andLTC3405A-1.8 are high efficiencymonolithic synchronous buck regu-lators specifically designed to workwith ceramic input and output ca-pacitors. Unlike the LTC3405, theinternal loop compensation of thesenew devices does not rely on the out-put capacitor ESR for stable operation.Ceramic output capacitors can beused freely for very low output rippleand small circuit size. Housed in tiny6-lead TSOT -23 packages, theLTC3405A, LTC3405A-1.5 andLTC3405A-1.8 can supply 300mA ofoutput current. Their high switchingfrequency (1.5MHz) allows the use ofvery small inductors and capacitors.The internal synchronous switch in-creases efficiency and eliminates theneed for an external Schottky diode.The LTC3405A provides adjustable

output voltage, while the LTC3405A-1.5 and LTC3405A-1.8 are fixed atoutputs of 1.5V and 1.8V respec-tively. The fixed output voltageversions eliminate the need for theoutput voltage setting resistors, fur-ther reducing the number of externalcomponents and saving space. A com-plete switching regulator solution canoccupy less than 0.06in2 of boardspace and require only three externalcomponents: an input capacitor, anoutput capacitor and an inductor.

The LTC3405A, LTC3405A-1.5 andLTC3405A-1.8 all use a constant fre-quency, current mode architecture toprovide excellent transient responseand line regulation. The supply volt-age ranges from 2.5V to 5.5V makingthem ideally suited for single Li-Ionbattery-powered applications. Thesupply current during operation isonly 20µA while maintaining the out-put voltage with no load (using BurstMode operation) and < 1µA in shut-down. This enables the regulators tomaintain better than 90% efficiencyover three decades of output loadcurrent. For noise-sensitive applica-tions, Burst Mode operation can bedisabled by connecting a MODE pinto VIN or driving it with a logic highsignal. This enables constant-frequency operation, which ismaintained at lower load currentstogether with lower output ripple. Ifthe load current is low enough, cycleskipping occurs to maintain regula-tion. In constant frequency mode, theefficiency is lower than Burst Modeoperation at light loads, but it is com-parable to Burst Mode operation whenthe output load current exceeds25mA.

All three devices can deliver 300mAin a tiny low profile 1mm height TSOT-23 package.

Rugged CAN TransceiverSurvives Loss of Ground andShorts to ±60VThe LT1796 is a rugged transceiverfor Controller Area Network (CAN)bus applications. The LT1796 canwithstand ±15kV ESD strikes andfaults up to ±60V. This makes it idealfor harsh environments, such as in-


NEW DEVICE CAMEOS

dustrial controls with 24V supplies,or heavy-duty truck applications. Inthese applications, the loss of groundconnection or cross-wiring faults canforce DC voltages in excess of 24V ineither polarity onto the bus pins. TheLT1796 can survive such faults with-out the need for external protectioncircuitry.

The LT1796 matches the industrystandard footprint in the SO-8 pack-age, including a combined slew ratecontrol/standby pin. In standby, thesupply current is reduced to 800µA.The slew rate control allows a maxi-mum data rate of 500kbps, or it canbe programmed for slower rates tominimize EMI and reduce reflectionsdue to long stubs or improper termi-nation.

LTC4211 Hot Swap ControllerProvides OvercurrentProtection and InrushCurrent Limiting Without anExternal Gate CapacitorThe LTC4211 Hot Swap™ controllerfeatures dual level overcurrent pro-tection and inrush current limitingwithout the need for an external gatecapacitor. The dual level overcurrentprotection is implemented using twocomparators, each with different

enhanced TSSOP16 exposedleadframe package provide high-power in a more compact solutionthan is possible with either dual con-trollers—at a much higher cost—or asingle controller and separately cho-sen MOSFET—a more complex designusing extra board space and designand assembly time. The B220ASchottky diodes have a low forwardvoltage rating for high efficiency anda small case size to further minimizeboard space. The ceramic input andoutput capacitors provide a tiny, low-cost solution with minimal outputripple.

The current-mode topology of theregulator provides stable response toload transients on both outputs—requiring only ceramic output

capacitors and a simple RC networklocated on the VC pin of theLT1765EFE. This is a space and costsaving advantage over a voltage-modecontroller topology, which would re-quire additional compensationcomponents to optimize load tran-sient response. Also, voltage-modecontrollers typically require electro-lytic or tantalum output capacitors,rather than extremely low ESR ce-ramic capacitors, to stabilize thecontrol loop and maintain good highfrequency response. Given the sameRMS current-handling requirement,electrolytic and tantalum capacitorstake much more space and createmuch more output voltage ripple thanthe equivalent ceramic. Overall, acurrent-mode step-down regulatorwith ceramic capacitors is simpler,

smaller, and less expensive than avoltage mode solution.

The swi tch current o f theLT1765EFE, which has a minimumrating of 3A, limits the maximumoutput current of the negative lineand positive line. In this topology, thenegative output current must be lessthan (and cannot equal) the positiveoutput current, or the output voltagewill drop out, so care must be takenwhen considering all possible load-transient conditions. The typicalmaximum negative output currentwith respect to the positive outputcurrent is shown in Figure 4. If cross-regulation is an issue with +5V outputcurrent greater than 1.0A and –5Vnegative output current less than5mA, a 1k preload resistor on the –5Voutput can improve regulation.

thresholds and response times, tomonitor the voltage across an exter-nal RSENSE resistor connected betweenthe VCC and SENSE pins. The slowcomparator helps to filter out noiseby tripping when the voltage acrossthe resistor exceeds 50mV for morethan 20µs. The fast comparator pro-tects the board against potentiallydamaging high energy voltage spikesand severe overloads—it trips whenthe voltage exceeds 150mV for morethan 300ns. During power-up, aninternal soft-start circuit graduallyramps up the GATE pin of an externalN-channel MOSFET to limit inrushcurrent to 50mV/RSENSE. The soft-start circuit does not require anexternal gate capacitor, which meansfaster turn-off times when an over-load occurs, and significant cost andspace savings.

The operation of the LTC4211 issequenced in two timing cycles, withthe duration of both cycles set by anexternal capacitor connected from theTIMER pin to ground. The first timingcycle—the plug-in cycle—providesenough time for a solid connection tobe made to the backplane and for thepower supply to settle. The secondtiming cycle determines the durationof the soft-start cycle. The ON pin ofthe LTC4211 must be taken high for

the first timing cycle to be activated.As the GATE voltage ramps up duringthe second timing cycle, the FB pinmonitors the load side voltage via anexternal resistor divider and forcesthe RESET open drain pin low untilthe voltage rises above some nominalvalue.

The LTC4211 operates from 2.5Vto 16.5V and is available in the SO-8,MSOP-8 and MSOP-10 packages. It isspecified for both commercial andindustrial temperature ranges. The8-pin versions are pin compatible withthe LTC1422, allowing for easy up-grades. The 10-pin versions addFILTER and FAULT pins. The 20µsresponse time of the slow comparatorcan be increased by connecting anexternal capacitor from the FILTERpin to ground. The FILTER pin canalso used in conjunction with an ex-ternal Zener diode to add overvoltageprotection. The FAULT pin is an opendrain output which is normally pulledhigh by an external pull-up resistor.It serves as a status indication andpulls low when the LTC4211 is latchedoff by an overcurrent fault. It can alsobe tied to the ON pin for auto-retryapplications in which the LTC4211tries to reconnect the load automati-cally after being latched off.



1990 Linear Databook, Vol I —This 1440 page collec-tion of data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conver-sion and interface products (bipolar and CMOS), in bothcommercial and military grades. The catalog featureswell over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 page supple-ment to the 1990 Linear Databook includes all productsintroduced in 1991 and 1992. $10.00

1994 Linear Databook, Vol III —This 1826 page supple-ment to the 1990 and 1992 Linear Databooks includesall products introduced since 1992. $10.00

1995 Linear Databook, Vol IV —This 1152 page supple-ment to the 1990, 1992 and 1994 Linear Databooksincludes all products introduced since 1994. $10.00

1996 Linear Databook, Vol V —This 1152 page supple-ment to the 1990, 1992, 1994 and 1995 Linear Databooksincludes all products introduced since 1995. $10.00

1997 Linear Databook, Vol VI —This 1360 page supple-ment to the 1990, 1992, 1994, 1995 and 1996 LinearDatabooks includes all products introduced since 1996.

$10.00

1999 Linear Databook, Vol VII — This 1968 pagesupplement to the 1990, 1992, 1994, 1995, 1996 and1997 Linear Databooks includes all products introducedsince 1997. $10.00

1990 Linear Applications Handbook, Volume I —928 pages full of application ideas covered in depth by40 Application Notes and 33 Design Notes. This catalogcovers a broad range of “real world” linear circuitry. Inaddition to detailed, systems-oriented circuits, this hand-book contains broad tutorial content together with liberaluse of schematics and scope photography. A specialfeature in this edition includes a 22-page section onSPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —Continues the stream of “real world” linear circuitryinitiated by the 1990 Handbook. Similar in scope to the1990 edition, the new book covers Application Notes 40through 54 and Design Notes 33 through 69. Refer-ences and articles from non-LTC publications that wehave found useful are also included. $20.00

1997 Linear Applications Handbook, Volume III —This 976 page handbook includes Application Notes 55through 69 and Design Notes 70 through 144. Subjectsinclude switching regulators, measurement and controlcircuits, filters, video designs, interface, data convert-ers, power products, battery chargers and CCFL inverters.An extensive subject index references circuits in LTCdata sheets, design notes, application notes and LinearTechnology magazines. $20.00

FilterCAD™ 3.0 — FilterCAD 3.0 is a computer aideddesign program for creating filters with LinearTechnology’s filter ICs. Filter CAD is designed to helpusers without special expertise in filter design to designgood filters with a minimum of effort. It can also helpexperienced filter designers achieve better results byplaying “what if” with the configuration and values ofvarious components and observing the results. WithFCAD, you can design lowpass, highpass, bandpass ornotch filters with a variety of responses, includingButterworth, Bessel, Chebychev, elliptic and minimumQ elliptic, plus custom responses. Download atwww.linear.com

SPICE Macromodel Disk — This IBM-PC (or compat-ible) high density diskette contains the library of LTC opamp SPICE macromodels. The models can be used withany version of SPICE for general analog circuit simula-tions. The diskette also contains working circuit examplesusing the models and a demonstration copy of PSPICE™by MicroSim. Available at no charge

Noise Disk — This IBM-PC (or compatible) programallows the user to calculate circuit noise using LTC opamps, determine the best LTC op amp for a low noiseapplication, display the noise data for LTC op amps,calculate resistor noise and calculate noise using specsfor any op amp. Available at no charge

www.linear.comand the Linear DirectOnline StoreLTC Web Site — Customers can quickly and conve-niently find and retrieve the latest technical informationcovering the company’s products on LTC’s web site.Located at www.linear.com, the site allows searching ofdata sheets, application notes, design notes, LinearTechnology magazine issues and other LTC publica-tions. The LTC web site simplifies searches by providingthree separate search engines. The first is a quick searchfunction that provides a complete list of all documenta-tion for a particular word or part number. There is alsoa product function tree that lists all products in a givenproduct family. The most powerful, though, is the para-metric search engine. It allows engineers to specify keyparameters and specifications that satisfy their designrequirements. Other areas within the site include a salesoffice directory, press releases, financial information,quality assurance documentation, and general corpo-rate information.

Linear Direct Online Store — As of May 1, 2002 theLinear Direct Online Store will be temporarily underreconstruction for approximately four to six weeks. Topurchase LTC products during this time, please contactyour local sales office or distributor.

DESIGN TOOLS

Acrobat is a trademark of Adobe Systems, Inc.; Windowsis a registered trademark of Microsoft Corp.; PSPICE is atrademark of MicroSim Corp.

Brochures and SoftwarePower Management Solutions Brochure — This 96page collection of circuits contains real-life solutions forcommon power supply design problems. There are over70 circuits, including descriptions, graphs and perfor-mance specifications. Topics covered include batterychargers, desktop PC power supplies, notebook PCpower supplies, portable electronics power supplies,distributed power supplies, telecommunications andisolated power supplies, off-line power supplies andpower management circuits. Selection guides are pro-vided for each section and a variety of helpful designtools are also listed for quick reference.

Available at no charge

Data Conversion Solutions Brochure — This 88 pagecollection of data conversion circuits, products andselection guides serves as excellent reference for thedata acquisition system designer. Over 40 products areshowcased, solving problems in low power, small sizeand high performance data conversion applications—with performance graphs and specifications. Topicscovered include delta-sigma ADCs, low power and highspeed ADCs and low power and high speed DACs. Acomplete glossary defines data conversionspecifications; a list of selected application and designnotes is also included. Available at no charge

Telecommunications Solutions Brochure —This 76page collection of application circuits and selectionguides covers a wide variety of products targeted fortelecommunications. Circuits solve real life problemsfor central office switching, cellular phones, high speedmodems, basestation, plus special sections covering–48V and Hot SwapTM applications. Many applicationshighlight new products such as Hot Swap controllers,power products, high speed amplifiers, A/D converters,interface transceivers and filters. Includes a telecom-munications glossary, serial interface standards, protocolinformation and a complete list of key application notesand design notes. Available at no charge

SwitcherCAD™ III — LTC SwitcherCAD III is a fullyfunctional SPICE simulator with enhancements andmodels to ease the simulation of switching regulators.This SPICE is a high performance circuit simulator andintegrated waveform viewer, and also includes sche-matic capture. Our enhancements to SPICE result inmuch faster simulation of switching regulators than ispossible with normal SPICE simulators. SwitcherCADIII includes SPICE, macromodels for 80% of LTC’sswitching regulators and over 200 op amp models. Italso includes models of resistors, transistors and MOS-FETs. With this SPICE simulator, most switchingregulator waveforms can be viewed in a few minutes ona high performance PC. Circuits using op amps andtransistors can also be easily simulated. Downloadat www.linear.com

Databooks andApplications Handbooks

DESIGN TOOLS

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LINEAR TECHNOLOGYLINEAR TECHNOLOGY · Linear Technology Magazine • May 2002 3 LTC1733, continued...

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