15. J.H. Chen, X.G. Huang, and Z.J. Huang, Simple thin-film fiber optic

temperature sensor based on Fabry-Perot interference, Opt Eng 49

(2010), 4.

16. A.D. Kersey, M.A. Davis, H.J. Patrick, M. LeBlanc, K.P. Koo, C.G.

Askins, and E.J. Friebele, Fiber grating sensors, J Lightwave Tech-

nol 15 (1997), 1442–1463.

17. P. Biswas, S. Bandyopadhyay, K. Kesavan, S. Parivallal, B.A.

Sundaram, K. Ravisankar, and K. Dasgupta, Investigation on pack-

ages of fiber Bragg grating for use as embeddable strain sensor in

concrete structure, Sens Actuators A 157 (2010), 77–83.

18. X.W. Wu, H.C. Xue, H.Y. Meng, W. Shen, and W. Wang, Simulta-

neous measurement of temperature and strain by combining a fiber

Bragg grating and the pigtail fiber covered with epoxy resin, Rev

Sci Instrum 82 (2011), 064904.

VC 2013 Wiley Periodicals, Inc.

DESIGN AND OPTIMIZATION OF NEWCOMPACT TUNABLE 2.4-GHz BANDPASS FILTER USING COUPLED k/2MICROSTRIP OPEN-LOOP RESONATORSAND MEMS-SWITCH TECHNIQUE

Ahmed Boutejdar,1 Abbas Omar,1 and Edmund Burte2

1 Chair of Microwave and Communication Engineering, Ottovon Guericke University Magdeburg, 39106 Magdeburg, Germany;Corresponding author:boutejdar69@gmail.com2 Chair of Semiconductor Technology, Otto von Guericke UniversityMagdeburg, 39106 Magdeburg, Germany

Received 4 March 2013

ABSTRACT: A new compact WLAN-band pass filters are proposedusing a half wavelength microstrip open-loop and k/2 microstrip resona-

tors based on direct-cross coupling method and micro electronicmechanical system (MEMS) technology. The filter is not only compact in

size due to the slow-wave effect, but also has multitransmission zeros atboth low and high rejection bands, as well as low insertion loss in thepass band. A demonstration filter has been designed and tested. Simula-

tion results almost agree with the measured results. Some discrepancybetween them can still be observed, which can be attributed to the unex-

pected tolerance of fabrication. Due to the simplicity of the design andits fabrication, it shows that this type of band pass filter has greatpotential for actual applications in communication and other systems.

By applying the MEMS technology, a switchable band pass filter basedon the quadruplet filter is able to be designed. VC 2013 Wiley Periodi-

cals, Inc. Microwave Opt Technol Lett 55:2444–2450, 2013; View this

article online at wileyonlinelibrary.com. DOI 10.1002/mop.27843

Key words: quadruplet filter; coupling matrix; Tunable band-pass filter;

MEMS-switch

1. INTRODUCTION

In the modern communication systems, a band pass filter is an

essential element. As the development of the communication

technology increases, the frequency spectrum becomes more and

more valuable. The frequency channels are much closed to each

other; therefore, a sharp pass band skirt is always required by fil-

ter design. Using conventional filter causes too much insertion

loss because of the high order. Introducing the finite transmission

zeros (TZs) is a good way to realize sharp skirt without high

order. The theory of cross-coupled filter is developed to generate

such TZs. Cameron has proposed a general method for the syn-

thesis of the coupling matrix [1]. He developed also a recursion

technique to synthesize the quasielliptic transmission function,

which is able to generate the prescribed TZs. Besides Cameron’s

synthesis method, the optimization method is also proved to be a

good option to extract the coupling matrix of cross-coupled fil-

ters. Amari improved Cameron’s recursion technique and made

the synthesis of transmission function simpler by programming

[2–5]. The optimization method is actually to minimize the cost

function, which contains the coupling matrix entries. In this the-

sis, the error function proposed by Hong [5] is used.

MEMS technology is in recent years quickly developed and

widely used in HF systems. “MEMS” is the abbreviation of

“Micro Electric Mechanic System,” which is usually fabricated in

unit of micrometer. A MEMS switch is able to serve as a fre-

quency switch. For example, the resistive MEMS switch is used

for 0.1–40 GHz applications, which has a high isolation around

220 to 260 dB by OFF-state and very low insertion loss around

20.1 to 20.2 dB by ON-state [6]. The most important part of the

MEMS switch is a moveable several micrometers thin metal

bridge, which can be pulled down by Coulomb force to switch on

the strip line and will be released to initial position to switch off

by restoring force of it self when the Coulomb force is cancelled.

This feature can be used to switch the length of microstrip resona-

tors. For application in WLAN system, a quadruplet filter (see Fig.

1) with working frequency at 2.4 GHz is designed and then this

quadruplet filter is combined with MEMS switches to achieve a

switchable filter, which can be switched either at working fre-

quency 2.4 GHz or 1.8 GHz. In this thesis, the RO4003C is used

as the substrate with relative dielectric constant 3.38 and thickness

0.813 mm. Strip line and ground are 0.035-mm thick copper.

2. OPEN-LOOP-RING RESONATOR

The open-loop-ring resonator is a basic component of the quad-

ruplet filter. We use here the square formed half-wavelength res-

onator (see Fig. 1). The a, w, and g indicate the square length,

strip width, and the gap width, respectively. The wavelength is

the guided wavelength at resonant frequency, which can be cal-

culated by [5]

kg05

k0ffiffiffiffiffiffiffiffiffiereð Þ

p 5c

f0

ffiffiffiffiffiffiffiffiffiereð Þ

p (1)

where k0, c denote the light wavelength and velocity in free

space, respectively, f0 is the resonant frequency, ere is the

Figure 1 Schematic view of the open-loop-ring resonator. [Color fig-

ure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

2444 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 DOI 10.1002/mop

effective dielectric constant. For resonant frequency at 2.4 GHz,

the guided wavelength can be calculated as 77.8 mm. The fields

at the end of the transmission line do not stop abruptly and result

in fringe fields, which can be modeled as a shunt capacitance Cp

between the open end and the metal ground or as an equivalent

transmission line with length Dl [5]. With the help of EM simula-

tion, the resonator dimensions are determined: a 5 10.3 mm,

w 5 1 mm, and g 5 1 mm. The length Dl can be calculated into

the length of the resonator and then the open loop ring resonator

is equivalent to a transmission line shunt circuited with a capaci-

tor Cg, which is caused by the open gap [5].

In Figure 2, Zc indicates the characteristic impedance of the

transmission line, c indicates the complex propagation constant,

and l is the equivalent length of the transmission line. The

admittance matrix of the lossy transmission line is given by [7]

Y15Y11 Y12

Y21 Y22

" #5

Yccoth clð Þ 2Yccsc h clð Þ

2Yccoth clð Þ Yccoth clð Þ

" #(2)

The admittance matrix of the capacitor is given by [7,8]

Y25jxCg 2jxCg

2jxCg jxCg

" #(3)

The input admittance can be derived as [7,8]

Yin ffi11jpDx=alx0

1=2Ycal(4)

where x 5 x0 1 Dx, is the attenuation constant, which has a

relation with complex propagation constant c 5 a 1 jb. The

open-loop-ring resonator can also be modeled as a lumped GLC

resonant circuit as demonstrated in Figure 3.

The input admittance of the conductance, inductance, capaci-

tor (GLC) circuit is:

Yin5G1jxC11

jxL5G12jDxC (5)

Comparing the Eqs. (4) and (5), we can get the values of

equivalent GLC elements.

G5akg0Yc

2(6)

C5pYc

2x0

(7)

L51

x20C

(8)

with the formulas for calculation of the attenuation constant [7–

12], we have G 5 1.4967.1024 S (Siemens), C 5 1.446 pF, and

L 5 3.016 nH. The responses of the open-loop-ring resonator

and the equivalent GLC circuit are compared in Figure 4.

3. THEORY AND DESIGN QUADRUPLET FILTER

In this section, a quadruplet filter is designed according to the

following specifications: center frequency f0 5 2426 MHz, pass

band 2382–2471 MHz, pass band return loss 20 dB, bandwidth

60 MHz, and fractional bandwidth (FBW) can be calculated as

0.0367 or 3.67%. The two symmetric finite TZs are chosen at

X 5 62, where X is the normalized frequency. By applying the

frequency transformation, the two TZs can be derived as

f1 5 2353 MHz and f2 5 2489 MHz. The filter structure is dem-

onstrated in Figure 5, where sij denotes the direct and cross cou-

plings between the ring resonators, whereas distance t denotes

the tap position of the feed line.

3.1. Transmission Function of Prototype FilterAccording to the design specifications given above, we can obtain

the transmission function of a prototype filter using quasielliptic

approximation [1,2]. For a lossless passive filter, we have

jS11 jXð Þj5 PN Xð ÞEN Xð Þ jS21 jXð Þj5 DN Xð Þ

eEN Xð Þ (9)

where DN(X) contains all the TZs, PN(X) contains all the return

loss parts. N indicates the filter order. Apply the power conser-

vation condition, that is, |S11(jX) |2 1 |S21(jX)|2 5 1. The

amplitude-squared transmission function is derived as

jS11 jXð Þj251

11e2 PN Xð ÞEN Xð Þ

� �25

1

11e2F2N Xð Þ (10)

where e is the ripple constant, which is in term of prescribed

pass band return loss LR in decibels as [5].

Figure 2 Equivalent transmission line model of open-loop-ring

resonator

Figure 3 Equivalent circuit of resonator with tapped feed line Figure 4 Comparison of EM and circuit simulation results

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 2445

e51ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi

100:1LR 21p (11)

The filter characteristic function is given by [1]

FN Xð Þ5coshXN

n51

cosh 21 xnð Þ" #

xn5

X21

Xn

12XXn

(12)

where Xn indicates the prescribed TZs. FN(X) can be trans-

formed in form of rational function by applying some identities

and arrangements given by Cameron [1].

FN Xð Þ5 PN Xð ÞDN Xð Þ5

PN Xð ÞYN

n5112

XXn

(13)

To compute the polynomial PN(X), Amari developed a recur-

sion process, which can be easily programmed [2].

PN11 Xð Þ52PN21 Xð Þ 12XXn

2

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi12

1

X2N11

sffiffiffiffiffiffiffiffiffiffiffiffiffiffi12

1

X2N

s

1PN Xð Þ X21

XN11

1 X21

XN

� �ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi12

1

X2N11

sffiffiffiffiffiffiffiffiffiffiffiffiffiffi12

1

X2N

s2666664

3777775

(14)

The first two recursion factors are given as:

P0 Xð Þ51; P1 Xð Þ5X21

X1

(15)

For our four-order filter with TZs at X 5 62, the following

polynomial can be derived with S 5 jX.

D Sð Þ50:25 S12jð Þ S22jð Þ

P Sð Þ56:9641 S10:9333jð Þ s10:406jð Þ

S20:9333jð Þ S20:406jð Þ

E Sð Þ56:9641 S11:181920:2459jð Þ

S10:585110:8018jð Þ S10:585120:8018jð Þ

(16)

Applying the formula in (9), the transmission and reflexion

response can be derived and they are plotted in Figure 6.

3.2. Extraction of Coupling MatrixFor the optimization method, the error function must be estab-

lished. The error function used here is [5]

EF X;Uð Þ5XI

i51

jSCAA21 Xi;Uð Þ2S21 Xið Þj2

1jSCAA11 Xi;Uð Þ2S11 Xið Þj2

24

35

(17)

where are the transmission and reflexion functions to approx-

imate the prototype functions derived in last section. They are

expressed as [5]

SCAA21 Xi;Uð Þ5 2ffiffiffiffiffiffiffiffiffiffiffiffi

qe1qe4p �A½ �21

41

SCAA11 Xi;Uð Þ512

2ffiffiffiffiffiffiq2

e1

q �A½ �21

11

(18)

where Xi is the sample frequency by optimization, U is a vector,

which contains all the variables to be optimized, that is, the

entries of the general immittance matrix[�A] of the filter.

Because of the symmetrical characteristic of the quadruplet

topology, we get the following equations qe15qe4,m12 5 m21 5 m34 5 m43, m23 5 m32, m14 5 m41 and there are no

couplings between either the Resonators 1 and 3 or 2 and 4,

therefore we have also m13 5 m32, m24 5 m42 5 0, where qe1, qe4

Figure 5 Schematic view of the designed quadruplet-coupled band

pass filter (BPF). [Color figure can be viewed in the online issue, which

is available at wileyonlinelibrary.com]

Figure 6 The simulation results of the proposed BPF using coupling

matrix method. [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

2446 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 DOI 10.1002/mop

are the external quality factors of the input- and output-

resonator, respectively, the mij indicates the normalized coupling

coefficient between resonator i and j The general immittance

matrix can be further simplified as [5,9]:

�A½ �5

1

qe1jX 0 0 0

0 jX 0 0

0 0 jX 0

0 0 01

qe1jX

2666666666664

3777777777775

2

0 m12 0 m41

m21 jX m23 0

0 m23 jX m34

m41 0 m34 0

2666666664

3777777775

(19)

SCAA21 X; qem12m14m23½ �

� �5

2

qe

�A½ �21

41

SCAA11 X; qem12m14m23½ �

� �512

2

qe

�A½ �21

11

(20)

So far the error function can be programmed and optimized

in Matlab. After 57 iterations by minimal error function value

172.66, we found the optimized values as:

qe50:626500358360753

m1250:82157551756936629

m14520:2059072292554327

m2350:7953758348795252

(21)

3.3. Determination of the Filter DimensionsAs the required external quality factor and coupling coefficients

are found, we can now determine the filter dimensions with the

help of EM simulation. Here, we use the CST Microwave Stu-

dioTM as the simulator [13]. The three coupling configurations

in the quadruplet filter will be simulated and from the simula-

tion results two characteristic frequencies can be read out

directly and with them the coupling coefficient can be calcu-

lated using [3–5]

Mij56f 22 2f 2

1

f 22 1f 2

1

� �(22)

where Mij is the coupling coefficient between resonator i and

j, f1 and f2 are the two characteristic frequencies. The external

quality factor can be calculated using [3–5]:

Qe5f 20

f2900 2f

1900

!(23)

where Qe is the scaled external quality factor, f0 is the reso-

nant frequency of the input/output resonator. f690� are the fre-

quencies where the phase S11 of is equal to 690�, which can be

obtained by simulating the input/output resonator. With the

equations [3–5]:

Qe5qe

FBW;Mij5mij:FBW (24)

The filter dimensions can be determined and listed in Table 1.

4. DESIGN AND OPTIMIZATION OF THE PROPOSED BANDPASS FILTER

Based on the design philosophy described in the previous sec-

tion, a quadrupled filter, as seen in Figure 7, has been simulated

using the physical geometries shown in Table 1. Microwave

Studio CSTTM has been continuously used for a full-wave simu-

lation. The band pass filter has a bandwidth of 3.7% and small

losses in both bands.

As shown in Figure 7, a compact quadruplet band pass filter

with a combination of two microstrip feed lines and four U res-

onators was designed. Figure 8 shows the coupling topology of

the filter without the dimensions of the structure. Black balls

represent resonators designated from 1 to 4 and lines symbolize

coupling between resonators. The solid line represents direct

coupling between adjacent resonators. The dashed line repre-

sents cross coupling [2] between nonadjacent Resonators 1 and

4 (m14, m13 and m24), whereas the coupling (m13 and m24)

between the Resonators 1 and 3 and Resonators 2 and 4 are

negligible. This is due to the sufficient large distance between

Figure 7 Three-dimensional view of the quadruplet band pass filter.

[Color figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

TABLE 1 Dimensions of the Proposed Filter

t S21 S14 S23

1.75 mm 1.6 mm 2.1 mm 2.2 mmFigure 8 The coupling topology of the proposed four-order band pass filter

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 2447

them. Therefore, this kind of coupling remains without signifi-

cant affect on the resulting frequency characteristic.

To calculate the optimal coupling distances between both

neighbored resonators on the top layer, a coupling matrix

method has been used. To obtain the coupling matrix of this

topology, the specifications of the filter are defined and then the

desired parameters are extracted using an optimization based

scheme [5]. The coupling coefficient and quality factor curves

[3] are then used to realize obtained coupling coefficients and

quality factors. To validate the design philosophy and design

parameters, the coupled band pass filter has been simulated on

an RO4003 substrate with a dielectric constant 3.38 and a height

0.813 mm. Figures 7 and 8 show the layout and the coupling

topology of the simulated filter. Simulated results of the inser-

tion loss and return loss are displayed in Figure 9. The filter

shows four frequency poles in pass band and two TZs at both

low and high rejection bands. The resonant frequency at 2.426

GHz is expected. Insertion loss is observed as 20.22 dB at 2.43

GHz, and the suppression levels are observed as 233.5 dB at

2.2 and at 3 GHz.

5. FABRICATION AND MEASUREMENTS

To verify the efficacy of this proposed topology, the quadruplet

BPF with an occupied area less than 20 3 15 mm2 is fabricated

Figure 11 Measured and simulated S-parameters of the proposed

WLAN band pass filter. [Color figure can be viewed in the online issue,

which is available at wileyonlinelibrary.com]

Figure 10 Photograph of the fabricated quadruplet WLAN band pass

filter. [Color figure can be viewed in the online issue, which is available

at wileyonlinelibrary.com]

Figure 9 The simulation results of the quadrupled band pass filter.

[Color figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 12 The layout of the Open-loop ring resonator with two

extended arms with l 5 3.4 mm. [Color figure can be viewed in the

online issue, which is available at wileyonlinelibrary.com]

Figure 13 Comparison between EM simulation results of extended-

and original resonator. [Color figure can be viewed in the online issue,

which is available at wileyonlinelibrary.com]

2448 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 DOI 10.1002/mop

on a RO4003 Substrate, with the same properties as in simula-

tion process, and measured using the network analyzer Wiltron

37347A as shown in Figure 10. Referring to the measured result

in Figure 11, the filter has a pass band response centered at

2.427 GHz with high out-of-band rejection, insertion-loss less

than 22.5 dB and return-loss greater than 15 dB. Two TZs at

the high and low rejection bands resulted in the realization of

high selectivity and isolation. As shown in Figure 11, a reasona-

ble agreement between theoretical and experimental results is

achieved. Slight deviation observed between the simulated and

measured results is attributed to fabrication tolerance in the

implementation of the filter. It is envisaged that the filter could

be further refined to improve its overall performance. The topol-

ogy and the results of the proposed structure make it a suitable

candidate for use in modern wireless communication systems.

6. DESIGN OF SWITCHABLE BAND PASS FILTER WITH MEMSSWITCH

The length of the resonator determines the resonant frequency.

If we add extra strips at the end of the gap to extend the resona-

tor to increase the equivalent inductance, the resonant frequency

will be changed. The resonator with extended arms is con-

structed and simulated in CST (see Fig. 12). The simulation

result comparing with the simulation result of the original reso-

nator is plotted in Figure 13 below. We can see that the reso-

nant frequency is obviously moved to lower frequency.

The idea is when we set MEMS switches between the gap ends

and the extended arms, we can control the resonant frequency by

controlling the switch states. The pass band of the quadruplet filter

using such switchable resonators is able to be switched too. We

combine the MEMS switches with the quadruplet filter with 6.8-

mm long extended arms (see Fig. 14) and simulate it, and the sim-

ulation result is shown in Figure 15. There are two available

switch situations. For Situation 1, all the switches are switched off

and for Situation 2, all the switches are switched on.

For Situation 1, as in Figure 15 depicted, the simulation

results show a narrow pass band filter with a FBW of 3.9%.

The attenuation is less than 235 dB at both reject bands and

two TZs have been appeared at 1.76 GHz and 1.88 GHz.

Although for Situation 2, the FBW is 5%, the attenuation is less

than 219 dB at both reject bands. The two TZs have been

regenerated at 2.32 and 2.44 GHz. The simulated insertion loss

is less than 0.1 dB for both switch situations.

7. CONCLUSION

In this work, a quadruplet band pass filter is realized and the

pass band and positions of TZs in measurement show a good

agreement with the simulation results. However, the pass band

insertion loss in measurement is too much, which reaches 3 dB.

The pass band return loss in measurement is not as good as that

in the simulation either. One reason should be that this filter is

too sensitive, a pretty slight variation of the couplings distance

in value of 0.01 mm could result in a big difference by simula-

tion or measurement, and hence the imprecision of coupling dis-

tance by design and fabrication could be a reason. Otherwise,

the unideal coupling between the SMA-adapter of the filter and

the coaxial cable of the network analyzer could also be a reason

of the high insertion loss. The designed switchable band pass

filter provides us two switchable pass bands at 1.8 and 2.4 GHz.

Because of the difficult fabrication of MEMS switches, the fab-

rication and measurement of this filter are not implemented.

ACKNOWLEDGMENTS

The authors would like to thank the German Research Founda-

tion (DFG) for financial support of this project. The authors are

Figure 14 Tunable band pass filter using MEMS switches. [Color fig-

ure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 15 Simulation results of the tunable band pass filter in differ-

ent situations. [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 2449

grateful to Mr. Harald Dempewolf and Mr. Abdulrahman Dar-

wish and Dennis Winkler for their supports and assistances for

their cooperation in laboratory.

REFERENCES

1. R.J. Cameron, General coupling matrix synthesis method for Cheby-

shev filtering functions, IEEE Trans Microwave Theory Tech 47

(1999).

2. S. Amari, Synthesis of cross-coupled resonator filters using an ana-

lytical gradient-based optimization technique, IEEE Trans Micro-

wave Theory Tech 48 (2000).

3. A. Boutejdar, A. Elsherbini, A. Balalem, J. Machac, and A. Omar,

Design of new DGS hairpain microstrip bandpass filter using cou-

pling matrix method, In: PIERS Progress in Electromagnetics

Research Symposium, Prague, Czech Republic, August 27–30, 2007,

pp. 261–265.

4. A. Boutejdar, A. Elsherbini, and A.S. Omar, Method for widening

the rejectband in lowpass/bandpass filters by employing coupled C-

shaped defected ground structure, IET Microwave Antenna Propag 2

(2008), 759–765.

5. J.S. Hong and M.J. Lancaster, Microstrip filters for RF/microwave appli-

cations, Copyright VC 2001 John Wiley & Sons, Inc., New York, NY,

2001, ISBNs: 0-471-38877-7 (Hardback); 0-471-22161-9 (Electronic).

6. G.M. Rebeiz, RF MEMS: Theory, design, and technology, Copyright

VC 2003 by John Wiley & Sons, Inc., Hoboken, NJ, 2003, ISBN: 0–

471-20169-3.

7. L.-H. Hsieh and K. Chang, Equivalent lumped elements G, L, C,

and unloaded Q’s of closed- and open-loop ring resonators, IEEE

Trans Microwave Theory Tech 50 (2002).

8. K. Chang, Microwave ring circuits and antennas, Wiley, New York,

1996.

9. S. Amari, U. Rosenberg, and J. Bornemann, Adaptive synthesis and

design of resonator filters with source/load-multiresonator coupling,

IEEE Trans Microwave Theory Tech 50 (2002), 1969–1978.

10. A. Boutejdar, S. Amari, and A. Omar, A novel compact J-

admittance inverter-coupled microstrip bandpass filter using

arrowhead-shape as defected ground structure (DGS), Microwave

Opt Technol Lett 52 (2010), 34–38.

11. R.A. Pucel, D.J. Mass�e, and C.P. Hartwig, Losses in microstrip,

IEEE Trans Microwave Theory Tech MTT-16 (1968), 342–350.

12. I.J. Bahl and R. Garg, Simple and accurate formulas for microstrip

with finite strip thickness, Proc IEEE 65 (1977), 1611–1612.

13. Microwave StudioTM Software, Version 12, CST Corporation, Wellesley

Hills, MA.

VC 2013 Wiley Periodicals, Inc.

STUDY OF LINK GAIN AND LOCKINGRANGE IN HYBRID MODE-LOCKEDLASER-BASED MILLIMETER-WAVERADIO-OVER-FIBER SYSTEM

Bilal A. Khawaja1 and Martin J. Cryan2

1 Department of Electronic and Power Engineering, PN-EngineeringCollege, National University of Sciences and Technology (NUST)Karachi, Pakistan; Corresponding author: khawaja.bilal@pnec.edu.pk2 Department of Electrical and Electronic Engineering, PhotonicsResearch Group, University of Bristol, Bristol, BS8 1UB, UnitedKingdom

Received 12 February 2013

ABSTRACT: An investigation of hybrid mode-locked lasers is per-formed to better understand their link gain and locking range behavior

under external RF injection for their use in millimeter wave radio-over-

fiber systems. The dependence of 40-GHz link gain on different parame-ters is initially identified and then the three main parameters of gain

and saturable absorber section biases and external RF injection signalpower are studied in detail. It is found that higher injection power leadsto lower link gain ranging from 219dB to 222dB, whereas lower injec-

tion power gives higher link gain ranging from 210dB to 219.9dB.This article also reports a hybrid mode-locking range of 90.97 MHzwhich is the widest reported so far for the technique used here. VC 2013

Wiley Periodicals, Inc. Microwave Opt Technol Lett 55:2450–2454,

2013; View this article online at wileyonlinelibrary.com. DOI 10.1002/

mop.27820

Key words: hybrid mode-locked lasers; millimetre wave; radio-over-fiber systems; link gain; locking range; RF injection signal

1. INTRODUCTION

The integration of fiber-optic and wireless networks form what is

often referred to as a radio-over-fiber (RoF) system [1], which has

become an important technology for the provision of untethered

access to broadband wireless communications especially operating

in millimetre-wave (mm-wave) frequency bands [2–6]. In these

bands, the problems of substantial propagation loss and line-of-

sight propagation can affect connections and synchronization

between different parts of the communication system [2,3]. The

RoF technique is considered as a promising solution to overcome

these problems because it takes the advantage of the low loss and

broadband bandwidth of optical fibers to distribute the mm-wave

carrier signal and data by modulating them onto an optical carrier.

At the remote end, the mm-wave signal is demodulated from the

optical carrier and radiated to the end-user [7], this approach can

dramatically reduce costs and simplify network architectures.

It has long been recognized and demonstrated [4,8,9] that

mode-locked lasers (MLLs) running under the hybrid mode-

locking (HML) regime offer an attractive and low-cost solution

for the generation and distribution of mm-wave signals in RoF

systems. State-of-the-art MLL devices [8,9] were extensively

characterized for HML and mm-wave phase shifting [10] for

applications in phased array antenna and smart-antenna beam

forming systems. Recently, wireless HML [8] for mm-wave

modulated wireless data transmission, self-oscillating mixer [9],

and photonic active integrated antenna [11] techniques were dem-

onstrated for use in next-generation 60-GHz RoF systems. Optical

fibers can be used to distribute mm-wave modulated radio signals

from these MLLs over long distances for gigabit per second Wi-

Fi [6,12,13]. Previously, extensive MLLs characterization showed

that “modulation enhancement” which can occur around 40 GHz

in the S21 amplitude response [8–10] leads to a “flat plateau”

which shows the locking range of the injection-locked MLL sys-

tem. However, the dependence of this modulation enhancement

or overall MLL link gain has not been studied previously.

A number of authors [4,14,15] have also reported mm-wave

modulated data transmission and microwave/mm-wave signal

generation using MLLs-based techniques. It has also been dem-

onstrated that by employing impedance-matching networks, RF

injection efficiency can be improved in MLLs; however, there

are no reports on the factors that control the MLL link gain. So,

it is felt that a thorough study is required to further understand

the factors which affect the link gain and the maximum achieva-

ble HML range in MLL-based mm-wave RoF systems.

This article extends some of the measurements setups per-

formed previously in [8–10] with additional MLL devices and

performs a study of link gain and locking range under the HML

regime which identifies the parameters that control the MLL

link gain. This article also reports for the first time, a HMLThis work was carried out at Photonics Research Group, University of Bristol, UK.

2450 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 10, October 2013 DOI 10.1002/mop