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536 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 65, NO. 2, FEBRUARY 2017

A Wideband Low-Power-Consumption22–32.5-GHz 0.18-μm BiCMOS Active

Balun-LNA With IM2 CancellationUsing a Transformer-CoupledCascode-Cascade Topology

Chadi Geha, Student Member, IEEE, Cam Nguyen, Fellow, IEEE, and Jose Silva-Martinez, Fellow, IEEE

Abstract— A low-power-consumption wideband 0.18-µmBiCMOS active balun-low noise amplifier (LNA) with linearityimprovement technique for millimeter-wave applicationsis proposed. The linearity technique utilizes constantGm transconductance structure with the second-orderintermodulation (IM2) cancellation that provides robustness toinput and output variations. The constant Gm is establishedwith equal emitters’ area ratios and proper base-emitter biasingvoltage, thus improving linearity. Furthermore, power savingis achieved using inductive coupling boosting the overall Gmstructure and reducing the current consumption for the auxiliarygain stage. The measured balun-LNA’s power gain between theinput and two outputs is 15.4 and 15.6 dB with input returnloss greater than 8.7 dB. The gain and phase mismatches areless than 1.8 dB and 12°, respectively. The balun-LNA noisefigures between the input and two outputs are less than 5.5and 6 dB at 32.5 GHz. The measured input points [referred1-dB gain compressions (Pin1dB’s), input referred third-orderintercept IIP3’s] and the input referred second-order interceptpoints (IIP2’s) are more than −14.6, −5.7, and 42 dBm across22–32.5 GHz, respectively, and the total power consumption isless than 9 mW drawn from 1.8 V power supply.

Index Terms— Active balun, balun-LNA, BiCMOS, CMOS, lownoise amplifier (LNA), radio-frequency integrated circuit (RFIC).

I. INTRODUCTION

RECENT Federal Communications Commission regula-tions have freed up some unlicensed millimeter-wave

(mm-wave) frequencies [1], [2]. Such regulations stem fromlower overcrowded spectrums and the increasing demandof users for high data rate wireless communications andradar sensors. Receivers targeting microwave and mm-waveapplications based on the wireless metropolitan area networkstandards ranging from 10 to 66 GHz, ultrawideband radar

Manuscript received January 23, 2016; revised July 4, 2016 andSeptember 24, 2016; accepted October 2, 2016. Date of publicationDecember 2, 2016; date of current version February 8, 2017. This paperwas made possible by NPRP Grant # 6-241-2-102 from the Qatar NationalResearch Fund (a member of Qatar Foundation). The statements made hereinare solely the responsibility of the authors.

The authors are with the Department of Electrical and Computer Engi-neering, Texas A&M University, College Station, TX 77843 USA (e-mail:[email protected]; [email protected]).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2016.2623778

vehicular sensor from 22 to 29 GHz, military radar forunmanned aerial vehicles from 35 to 37 GHz [3], and soon are essential to achieve the user end demands. This fre-quency spectrum allocation still encounters adjacent channelcoexistence, similar to lower frequency spectrums, like radioastronomy at 23.6–24 GHz, industrial-scientific-medical at24.05–24.25 GHz, local multipoint-distribution system at31 GHz, and cloud radar at 35 GHz [4]. In fact, it presentsa dilemma for some sensitive frequency bands where over-lapping exists. The design of silicon-based radio-frequencyintegrated circuit (RFIC) receiver front ends at these frequen-cies for wideband performance with simultaneously high gainand high linearity is very challenging. The low-noise amplifier(LNA) plays a crucial role in achieving high gain and linearityover wide operating frequency ranges for these receivers.Active balun-LNAs are capable of providing differential out-puts from a single-ended input and are important componentin receivers. Various wideband active balun-LNAs on sili-con at low frequencies, which implement active and passivefeedback mechanisms to improve linearity, gain, and phasemismatches, have been reported [5], [6]. However, employingactive feedback comes at the expense of power and nonlin-earity rendering the harmonics cancellation ineffective [6].A linearization technique based on derivative superpositionand its improved derivative version tend to provide impres-sive input referred third-order intercept point (IIP3) [7], [8].The derivative superposition methods use auxiliary n/pMOSpath in weak inversion to cancel the third-order nonlinearcurrent of the main transconductance gain-stage path, thusenhancing IIP3. Nonetheless, this improvement is subjectto deter the second intermodulation product (IP2) due tononlinear cross terms between the two paths [7]. Further-more, current-mode balun-LNA-based common-gate common-source structures with bias control and output conductancekept constant show optimum behavior for both noise andlinearity [9], [10]. Such a constraint across wideband iscostly in terms of power consumption and subject to process,voltage, and temperature variations. Anotherapproach is mak-ing the third intermodulation (IM3) cancellation independentof frequency in bipolar junction transistor (BJT) [11]–[13].A second-harmonic control with fully differential mode

0018-9480 © 2016 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

GEHA et al.: WIDEBAND LOW-POWER-CONSUMPTION 22–32.5-GHz 0.18-μm BICMOS ACTIVE BALUN-LNA 537

configuration using BJT devices facilitates frequency-independent IM3 cancellation [11]. In [12] and [13], IM3 can-cellation happens due to current hyperbolic tangent behaviorfrom dual gated BJT devices in differential and pseudodif-ferential modes added to the output. However, the cost isdoubled in noise and power consumption. All of these tech-niques were implemented in designs operating below 2.4 GHz.A 20-GHz balun-LNA using 0.25-μm SiGe BiCMOS tech-nology was reported in [14]. This balun-LNA consists ofa common-emitter (CE) gain stage followed by a single-to-differential output buffer stage using a CE common-base(CE-CB) structure with ac current source. This designsuffers from very high phase and gain mismatches,thus limiting the bandwidth. In [17], a dc-50-GHzbalun-LNA based on distributed (CB-CE) structure usingtransmission line for phase correction in 90-nm CMOS wasdeveloped. In addition, a dc-50-GHz balun using (CE-CC)design approach was shown in [18] using 0.18-μm SiGeBiCMOS technology. A single transistor with feedback net-works utilized as balun based on signal-splitter topology (CE-CC) operates from dc-70 GHz using 0.35-μm SiGe BiCMOStechnology [19]. Furthermore, [23] exhibited a balun-LNAstructure with two parallel balanced differential amplifiersoperating from 60 to 67 GHz using 90-nm CMOS designprocess. Finally, a dc-60-GHz balun-LNA with distributedcommon-gate common-source structure using 65-nm CMOStechnology was reported in [24]. These works show a tradeoffbetween linearity, power consumption, and gain.

In this paper, a 0.18-μm SiGe BiCMOS 22–32.5-GHz activebalun-LNA with high linearity and low power consumptionis presented. The linearity improvement is attained using anew linearity technique based on a constant Gm-cell transcon-ductance that forms the balun-LNA structure. The constantGm-cell transconductance is established through equal emit-ters’ area ratios of the balun-LNA. The constant small-signalGm-cell transconductance remains independent of input andoutput variations under large-signal behavior and provides thesecond-order intermodulation (IM2) cancellation, resulting inimproved linearity. The low power consumption is due inpart to the coupled inductors used between cascaded stages.The balun-LNA targets multistandard multichannel receivers’applications ranging from 22 to 32.5 GHz that require highlinearity. Many microwave and mm-wave applications not onlycoexist, but also overlap each other on the same frequencyspectrum, making the linearity the bottle neck for the receiver’sdynamic range.

This paper is organized as follows. Section II dis-cusses the proposed balun-LNA architecture. Section IIIdescribes the design and implementation of coupled induc-tors model. Section IV provides the simulation and experi-mental results, and the conclusion remarks are presented inSection V.

II. PROPOSED BALUN-LNA ARCHITECTURE

Fig. 1 shows the schematic of the 22–32.5-GHz (single-to-differential) wideband active balun-LNA with high-gain, high-linearity, and low power consumption. Table I shows all the

Fig. 1. Schematic of the proposed balun-LNA.

TABLE I

COMPONENT PARAMETERS FOR THE PROPOSED BALUN-LNA

design parameters of the component in Fig. 1. The proposedbalun-LNA architecture consists of a main transconductancegm gain stage, Q1, coupled to an auxiliary gain path, Q2,through a transformer. The coupled transformer increases thesignal swing at the input of the second stage, thus boostingthe Gm transconductance, hence gain, and reducing the powerconsumption. The composite Gm cell defined by transistors I1,Q2, and Q3 plays a major role in improving the linearizationof the structure. The stipulated total Gm stays constant evenin the presence of variations in gm1 of Q1 and gm2 of Q2 dueto high input power. As the collector currents of transistorsQ1and Q2 vary from their quiescent bias under large voltageswing, the gm’s dependency on equal emitters’ area (Ae) ratioskeeps the overall Gm-cell constant. The overall Gm’s constantand frequency-independent characteristic behavior with IM2cancellation results in linearity enhancement. A simple wide-band input matching network is established using inductors Lb

and Le1 similar to [15]. The effect of the coupling transformers(Le1 and Lb2) on the input matching is considered thoroughlyin Section II-A. Inductive shunt peaking is used at the outputloads to extend the matching bandwidth of the balun-LNA.Finally, the noise due to the cascode transistor Q3 is reduced

538 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 65, NO. 2, FEBRUARY 2017

Fig. 2. Small signal model of the balun-LNA’s input impedance. gm is thesmall signal transconductance of Q1. Req2 is defined as ωT Le2 of Q2. Ipand Is are the primary and secondary currents of the transformer.

by adding an inductor Lm to resonate away the parasiticcapacitance at the emitter, thus reducing the output noise.All of these design techniques are implemented to designthe 22–32.5-GHz active balun-LNA.

A. Input Matching

Fig. 2 shows the small-signal input impedance of the balun-LNA derived from its schematic in Fig. 1. To keep theanalysis simple, the input impedance of the balun-LNA issplit into two sections Z B and Z ′

B , which represent the inputimpedances looking into the respective networks. Under theperfect matching condition, Z B = Z ′∗

B . Z B forms a pi-networkwith wideband matching characteristics, whose quality factor(Q) reduces due to the loading of the network representedby Z ′

B . For the ac coupled transformer (Le1 and Lb2) in Z ′B ,

the coupling coefficient K and the number of turn n can causethe optimum matching point to shift, yet keeping widebandimpedance matched to the input port. To study this effect, anexpression for the complex conjugate impedance Z ′∗

B is derived.Z ′∗

B is found using the small-signal model in Fig. 2 whereasthe adapted transformer model is similar to that in [16].Applying Kirchhoff current law (KCL) at nodes E1, C1,and B2, where M is the mutual inductance, K =(M/(L P LS)1/2) is the coupling coefficient, and n =(LS/L P )1/2 is the turn ratio of the ac coupled transformer,can lead to Z ′∗

B . Cpad is defined as the parasitic capacitancedue to RF pad on chip. Cbe, Cbe2, and Cbe3 are the parasiticcapacitances at the base-emitter junctions of transistors Q1,Q2, and Q3, respectively. In addition, Cbc and Cp2 are thecapacitances at the base-collector junction of transistors Q1and Q2. The KCL equations yield, after several manipulations

V ′B(s) = i ′

B(s)

(1 + 1

sCbe(1 + sgm Le1)

)− Ms Is (1)

where V ′B(s) is the base voltage looking into Z ′

B network port,and i ′

B(s) is its current defined as

i ′B(s) = sCbevbe. (2)

The secondary current Is of (1) can be derived as

Is = i ′B(s)

[gm Z1 − sM (sCbe + gm)

sCbe (Z1 − sLb2 − Z2)

](3)

Fig. 3. Comparison of the magnitudes of Z ′B with and without trans-

former (Xmr).

where

Z1 = sLm + 1

gm3 + sCbe3(4)

Z2 =(

1sCbe2

+ sLe2 + Req2

)

1 + 1/sCp2

(1

sCbe2+ sLe2 + Req2

) . (5)

Substituting Is into V ′B(s) and taking the ratio between (1)

and (2) gives

Z ′B(s) = 1 + 1

sCbe

×[

1 + sLe1

(gm − s (K n) gm Z1

Z1 − sLb2 − Z2

+ s2(K n)2sLe1 (gm + sCbe)

Z1 − sLb2 − Z2

)](6)

where Z ′B(s) shows that any changes in the coupling coef-

ficient K or the number of turn ratio n for the coupledtransformer can affect the poles and zeros alike, thus causingthe matching to shift into higher frequency, yet maintainingthe wideband characteristics due to poles-zeros cancellationeffect. Fig. 3 shows the schematic simulation and that fromthe analytical equation (6) for the magnitude of Z ′

B(s) withand without the transformer. It is clear that the widebandmatching characteristic is maintained with only small varia-tions of less than 4.3 � at 20 GHz in the worst case dueto unavoidable high-frequency effects on the small signalmodel.

B. Linearity

Fig. 4 shows the linearity model analysis for the conven-tional CE gm stage as well the proposed balun-LNA Gmstructure including the effect of the transformer. Using Taylorseries expansion approximation, the output collector currentfor the CE stage shown in Fig. 4(a) is given by

iC ∼= gm

⎡⎣ ∞∑

q=1,2,...

VT

q!(

vin

VT

)q⎤⎦ (7)

where gm = IQ1/VT , with IQ1 being the quiescent current ofQ1 and VT being the thermal voltage, is the voltage to current

GEHA et al.: WIDEBAND LOW-POWER-CONSUMPTION 22–32.5-GHz 0.18-μm BICMOS ACTIVE BALUN-LNA 539

Fig. 4. Linearity model analysis. (a) Conventional CE stage. (b) ProposedGm stage.

conversion also known as the small signal transconductancegm and vinis the input voltage. From (7), taking the qthorder derivatives of gm with respect to vin encompasses allnonlinearities for the CE stage.

Assumingvin = Va cos ωt and taking the ratio between thesecond and the fundamental harmonic amplitude in a CE stagegives the second-order harmonic distortion as

HD2 = 1

4

[Va

VT

]. (8)

The collector currents in the proposed Gm stage for thebalun-LNA, as shown in Fig. 4(b), can be derived using (7)as

iC1 ∼= IQ1

(vin

VT+ 1

2!(

vin

VT

)2

+ 1

3!(

vin

VT

)3

+ · · ·)

iC2 ∼= IQ2

(v2(1 + nK )

VT+ 1

2!(

v2(1 + nK )

VT

)2

+ 1

3!(

v2(1 + nK )

VT

)3

+ · · ·)

iC3 ∼= IQ3

(−v2

VT+ 1

2!(−v2

VT

)2

+ 1

3!(−v2

VT

)3

+ · · ·)

.

(9)

Using (9), we find the differential output current iOut =iC3 − iC2 with respect to the input voltage vin, assumingiC1 = iC3 and using the fact that −v2/vin = −gm1/gm3 =−Ae1/Ae3, where Ae1 and Ae3 represent the emitter area forQ1 and Q3, respectively, as

iOut =

⎧⎪⎪⎪⎪⎨⎪⎪⎪⎪⎩

IQ3

[−(Ae1/Ae3)vinVT

] [1 + IQ2(1+nK )

IQ3

]

+ IQ32

[(Ae1/Ae3)vin

VT

]2 [1 − IQ2(1+nK )2

IQ3

]

+ IQ36

[−(Ae1/Ae3)vinVT

]3 [1 + IQ2(1+nK )3

IQ3

]+ · · ·

⎫⎪⎪⎪⎪⎬⎪⎪⎪⎪⎭

.

(10)

Substituting vin = Va cos ωt into (10) results in

iOut

=

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

[IQ3

(−(Ae1/Ae3)VaVT

)]

×⎡⎢⎣(

1 + IQ2(1+nK )IQ3

)−

18

((Ae1/Ae3)Va

VT

)2 (1 + IQ2(1+nK )3

IQ3

)⎤⎥⎦ cos ωt

+ IQ34

((Ae1/Ae3)Va

VT

)2(1 + cos 2ωt)

(1 − IQ2(1+nK )2

IQ3

)

+ IQ324

(−(Ae1/Ae3)VaVT

)3(cos 3ωt)

×(

1 + IQ2(1+nK )3

IQ3

)+ · · ·

⎫⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎬⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎭

.

(11)

From (11), considering the ratios between the second andthe fundamental amplitude harmonics as well between thethird and the fundamental amplitude harmonics for the pro-posed Gm stage gives HD2,Gm and HD3,Gm, respectively, as(12) and (13), shown at the bottom of the next page.

As can be seen from (12), the cancellation of the non-linearity factor generated due to HD2,Gm is obtained underthe condition IQ2(1 + nK )2 = IQ3, which means gm2 =(gm3/(1 + nK )2) and, in turn, VBE3 ≈ VBE2 and Ae2 = Ae3.Hence, the overall Gm stays constant even in the presenceof variations in gm1 and gm2 due to large input voltagesignal. As the collector currents differ from their quiescent biasunder large input power, the gm’s dependency on the emitterarea ratios keeps the overall Gm constant. This large signalconstant gm characteristic results in linearity improvement.As HD3,Gm from (13) cannot be canceled, (13) dictates thelinearity limitation for this proposed architecture. However,there is a clear tradeoff between gain and linearity for thisbalun-LNA architecture. Keeping the aspect ratios Ae2 =Ae3 = Ae4 and gm2 = (gm3/(1 + nK )2) = gm4 between Q2,Q3, and Q4 maximizes the linearity at the expense of gaindue to Gm = (gm1 + ((gm1/gm3) + nK ) gm2). In addition,the gm2 transconductance increases due to the transformer’sproduct nK , which helps boost the gain for less dc current.However, given the transformer inductors’ sizes and the lim-ited nK value, the linearity degradation is very small, as shownin Fig. 5, which shows the simulation results of the inputreferred 1-dB gain compression (Pin1dB) for a cascode LNA,the proposed balun-LNA with and without transformer, andthe analytical derivation based on (13). All circuits consume6.4-mA current from a 1.8 V supply and achieve 16-dB powergain. The Pin1dB values for the regular cascode LNA, theproposed balun-LNA with and without transformer in outputdifferential modes, and the analytically theoretical simulationsare −15.8, −13.37, −13.26, and −13.34 dBm, respectively.The theoretical analysis matches well the Pin1dB simulationof the proposed structure. The linearity improvement of thebalun-LNA with transformer as compared with the cascodeLNA is better than 2.43 dB.

C. Noise Analysis

The noise of the proposed balun-LNA is dominated bythe input stage, including the matching network and its

540 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 65, NO. 2, FEBRUARY 2017

Fig. 5. Comparison curves for (a) cascode LNA, (b) balun-LNA withtransformer (Xmr), (c) balun-LNA without transformer (Xmr), and (d) theo-retical simulation.

Fig. 6. Noise sources model of the proposed balun-LNA.

auxiliary path. Fig. 6 shows the circuit’s main noise sourcesfor the proposed balun-LNA. The noise sources include baseand collector noise currents of Q1 and Q2. Noise due tothe parasitic base resistances Rbx and Rbx2 of Q1 and Q2,respectively, and noise due to the losses of Lb and RLb, andcoupling transformers Le1 and Lb2, and RLe1 and RLb2 areconsidered in the noise model. The noise due to the cascodetransistor Q3 is considerably reduced due to inductor Lm

rendering the degenerated impedance high at resonance, thusmaking its noise contribution negligible [15]. Furthermore,noise in the auxiliary path due to cascode transistor Q4 isneglected due to multicascaded transconductance gain stagesand, as a result, all cascode transistors are neglected in thefollowing analysis.

The equivalent input-referred noise due to the base andcollector current shot noise of Q1 and Q2, and its baseparasitic resistance Rbx2 are given by (A8)–(A12), shownat the bottom of page 546. According to (A8) and (A9)from the Appendix, the input referred noises of Q1 increaseproportionally with Lb inductor’s loss. This is because thesignal-to-noise ratio (SNR) between the input and the emitter-base junction is inversely proportional to Lb. It is clear thatthere is a tradeoff between the input matching requirementfor power transfer and the noise figure (NF) for this balun-LNA structure. However, (A8) and (A9) reflect the effect ofthe coupling transformer on the emitter impedance Ze of Q1.A higher Ze helps improve the collector current noise at theexpense of lower (SNR) at the emitter-base junction. Similarly,(A10)–(A12) show an increase in the SNR at the base-emitterjunction of Q2 raising the voltage gain through the couplingtransformer by (nk) factor. The collector shot noise of Q2 andits parasitic base resistance noise Rbx2 are improved by thesame factor.

The total input referred voltage noise due to Q1 and Q2,v2

ni,Q1,2, normalized to the noise voltage source impedance is

given by

v2ni,Q1,2

4kTRS� f≈ �1 (ω)

gm1+ (�2 (ω)) gm1 + �3 (ω)[

gm1gm3

+ nK]

+ (�4 (ω))

[gm1

gm3+ nK

]gm2+ �5 (ω)[

gm1gm3

+ nK]

gm2

(14)

where �1 (ω)−�5 (ω) are given by (A13)–(A17), shown at thebottom of page 546. This result shows that the collector currentshot noise of Q1 and Q2 can be improved by increasing gm1,gm2, and transformer’s product nK, respectively. However,such improvement comes at the expense of degrading the basecurrent shot noise. Hence, there is an optimum value for gm1and gm2 to minimize the total input-referred noise voltagedue to Q1 and Q2. Differentiating the first two terms andthe last two terms of (14) with respect to gm1 and gm2,respectively, and equating the resultant expressions to zeroresult in gm1,optand gm2,opt, given by

gm1,opt =√

�1 (ω)

�2 (ω)(15)

gm2,opt =√

�5 (ω)

�4 (ω)

1[gm1gm3

+ nk] . (16)

HD2,Gm =

∣∣∣∣∣∣∣∣

⎧⎪⎪⎨⎪⎪⎩

( 14

) [ (Ae1/Ae3)VaVT

] [1 − IQ2(1+nK )2

IQ3

][( 1

8

) ( (Ae1/Ae3)VaVT

)2] [

1 + IQ2(1+nK )3

IQ3

]−[1 + IQ2(1+nK )

IQ3

]⎫⎪⎪⎬⎪⎪⎭

∣∣∣∣∣∣∣∣(12)

HD3,Gm =( 1

24

) [ (Ae1/Ae3)VaVT

]2 [1 + IQ2(1+nK )3

IQ3

]{[

1 + IQ2(1+nK )IQ3

]− ( 1

8

) [ (Ae1/Ae3)VaVT

]2 [1 + IQ2(1+nK )3

IQ3

]} (13)

GEHA et al.: WIDEBAND LOW-POWER-CONSUMPTION 22–32.5-GHz 0.18-μm BICMOS ACTIVE BALUN-LNA 541

Fig. 7. NF for the differential output balun-LNA with ideal couplingcoefficient, K , and transformer multiple turns N .

The third term in (14) is due to the parasitic base resistancenoise, Rbx2, is limited by gm1,opt, and Ae2is the emitter area oftransistor Q2, and the transformer coupling factor (nK). Thetotal input referred NF of the proposed balun-LNA structureis given by

NFtot (ω)| ≈ 1 + RLb + Rbx

RS

(1 + ω2Cpad RS

)

+ 2√

�1 (ω) �2 (ω) + �3 (ω)[1

gm3

√�1(ω)�2(ω) + nk

]

+ 2√

�4 (ω) �5 (ω). (17)

Fig. 7 shows the analytical and NF simulations for thedifferential output of the balun-LNA with and without varioustransformer turns’ ratios. It is important to note that the impacton the minimum NF (NFmin) of different topologies of theproposed balun-LNA comes from the structure’s small signaltransconductance, Gm, variations due to its dependency onthe transformer turns’ ratios, particularly the transistor Q2.Furthermore, NFmin experiences a small effect due to thechanging behavior of the input matching network, where thetransformer turns’ ratios play a role, as seen in Section II-A.However, the impedance matching condition is maintainedacross the frequency of interest even for different N turns.Note that N = 1 is the only case scenario, where noisematch is established. The analytical result maps closely tothe simulator up to 35 GHz. From (17), it is clear that theSNR degradation between the source generator and the base-emitter junction capacitance is due to the matching inductanceloss Lb, RLb, the parasitic base resistance, Rbx, and padcapacitance, Cpad. In addition, the resistance 1/gm3 controlsthe tradeoff between the noise and linearity, where a smallerresistance improves the linearity at the expense of higher NFand lower gain. Furthermore, an increase in the turn ratioof the coupling transformer could improve the NF. However,the turn ratio cannot be increased randomly considering thecoupling transformer nonidealities [20]. Losses associated withparasitic resistances and capacitances at the base of Q2 mea-sure quadratically compared with the secondary inductanceof the transformer. Hence, the self-resonance frequency ofthe inductance suffers as well as the magnetic coupling, M ,reflecting higher noise. Ultimately, there are practical limits for

the voltage gain boosting effect and the optimal turn ratio n,thus achieving the lowest NF.

D. Stability and Power Efficiency

The effects of capacitors Cbc and Cp2 on both channelsare reduced due to the cascode structure. The added tran-sistors, Q3 and Q4, transform the input impedances of thedriving stages from negative impedances into a capacitiveone, and hence the stability is maintained. To illustrate thisconcept, we consider the input impedance looking into thebase of a simple CE structure loaded by an LC tank. Theinput impedance,Z in(ω), is a negative impedance derived asZ in(ω) ≈ (1/ jωCπ1)||(1/ jωCbc)||(−1/ω2CbcgmL), whereCπ1 and L are the base-emitter junction capacitance and loadinductance, respectively. To alleviate the negative-impedanceproblem, a cascode device is added to the CE structure. Theinput impedance, Z in(ω), then becomes purely capacitive asZ in(ω) ≈ (1/ jωCπ1)||(1/ jωCbc(1 + gm1/gm2)), where gm1and gm2 are the small-signal transconductances of the drivingand cascode transistors, respectively. Note the importance ofadding a cascode device to limit the miller gain effect on theparasitic capacitance Cbc to a factor of 2 for equal transconduc-tances (gm1 = gm2) with identical devices. However, there isa stability constraint for the cascode devices Q3 and Q4, wherethe input impedances looking into the emitters of Q3 and Q4are dominated by the parasitic base resistancer rbx3/4/β(s).At high frequencies, the ac gain factor, β( jw), is degraded andthe emitters’ input impedances start to increase with frequency,thus causing an inductive like behavior, which makes theactive balun-LNA structure prone to oscillation. To preventthis possible oscillation from happening, a large capacitanceCB is placed very close to the base of the cascode devices,as seen in Fig. 1, to absorb all inductances associated withthat node, including bonding wire and biasing interconnects.Furthermore, the transformer is designed in inverting config-uration to provide gain boosting without compromising thebalun-LNA stability.

The proposed balun-LNA structure having dual gm outputfrom a single-ended input combines the LNA characteristicwith the balun behavior into a single block. The invertingcoupling transformer boosts gm2 by (nK) factor. This topologyhas two properties: 1) it can further boosts the voltage gainat the base-emitter junction, thus reducing the dc bias point fora specific gain target, which means less dc power consumptionand 2) by controlling the coupling coefficient polarity, K,through proper layout of the stacked transformer, the voltagegain can be increased (with positive K ) or remains the samewith bandwidth extended (for negative K ).

E. Gain and Phase Balances

Fig. 8 shows a simplified schematic of the proposed balun-LNA, including the parasitic capacitances, used in the analysisof the gain and phase errors from node vx to output nodes vo1and vo2. The circuit between the RF input port to vx does notincur imbalance and hence is not considered in the analysis.The voltage gain transfer function from vx to the common-base

542 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 65, NO. 2, FEBRUARY 2017

Fig. 8. Simplified schematic of the active balun-LNA for the analysisof the gain and phase imbalance. C f , C f 2, and C p2 include all parasiticcapacitances at each particular node. Cin and C f are the total parasiticcapacitances at the emitter and collector node of Q3, respectively.

(noninverting) output vo1 is derived as

vo1

vx=

⎡⎢⎢⎢⎢⎣

sLd1

sLm· gm3

gm3/α + 1/sLm

· 1

1 + sCin

gm3/α+ 1/sLm

·(

1

1 + s2C f (Ld2+ Ld1)

)⎤⎥⎥⎥⎥⎦(18)

assuming rbx = 0 and ro = ∞ for all active devices. Thevoltage gain for the inverting path between vx and v02 can bederived as

vo2

vx

≈

⎡⎢⎢⎢⎢⎢⎢⎣

sLd3

Zs·

α(

sCμ2 − (1+nK )gm2(1+s Le2gm2(1/α+nK ))

)

s[Cπ2 +

(1 + (1+nK )gm2

gm4(1+s Le2gm2(1/α+nK ))

)Cμ2

] ·1(

1 + s(Cμ2+C2)gm4

/α

) · 1(1 + s2C f 2 (Ld3 + Ld4)

)

⎤⎥⎥⎥⎥⎥⎥⎦

(19)

where C2 is the parasitic capacitance at the collector terminalof Q2 and Zs = (1/sCc + sLb2) is the impedance of thenetwork in the signal path between vx and the base of Q2,with Cc being the coupling capacitor used as a mean for signalgain control in the auxiliary path as well as a neutralizationmechanism for gain and phase errors. The gain and phaseimbalance can be expressed, respectively, as

�v = 20 log10

×(√

1+ (ωZs(Cπ2+ 2Cμ2)+ ω2C f 2(Ld3 +Ld4))2

1 + (ω2C f (Ld1 + Ld2) − ω(Ld3/Le2)Cμ2)2

)

�φ = tan−1(ωZs(Cπ2 + 2Cμ2) + ω2C f 2(Ld3 + Ld4))

− tan−1(−ω(Ld3/Le2)Cμ2 + ω2C f (Ld1 + Ld2)) (20)

making use of(1 + (1 + nK ) gm2

gm4(1 + sLe2gm2 (1/α + nK )

))

≈ 2

due to identical Q2 and Q4 and the Q2’s small-signal transcon-ductance of

(1 + nK ) gm2(1 + sLe2gm2 (1/α + nK )

) ≈ 1

sLe2

at high frequencies. To obtain balanced gain and phasebetween the two signal paths, the poles and zeros of bothsignal paths’ transfer functions are set equal, which lead tothe following constraint for Cc:

Cc ≈ RGLmC f

(Cπ2+2Cμ2)C f 2− Lb2

(21)

where

RG = −gm3(1 + sLe2gm2

(1/α + nK

))(1 + nK ) gm2

and s2Cμ2 Le2 � 1.

III. TRANSFORMER AND INDUCTOR LAYOUT

The presence of the parasitic capacitors and resistive lossesgenerated from routing paths in integrated circuits causeslower quality factor in passive components, which could besignificant at mm-wave frequencies. To accurately accountfor such effects, all inductors and interconnects are simulatedusing electromagnetic (EM) simulator IE3D (Mentor Graph-ics, HyperLynx 3D EM) and the resultant S-parameters areimported into Cadence for circuit simulations. Inductors Ld1,Ld3, and Lb are designed using spiral inductor due to theirrelatively large inductances. However, a careful considerationis being assigned for the metal width trading off the resistiveloss, parasitic coupling to the substrate, quality factor, andinductors self-resonance frequencies. To guarantee inductorsbehaviors at mm-wave frequencies, it is important to achievethe quality factor peak beyond the frequency of interest.To reduce all type of losses, the top metal M6 is chosen for allinductors. Furthermore, inductors Lm , Le2, Ld2, and Ld4 andthe coupling transformers Le1 and Lb2 are all implementedusing microstrip transmission lines.

The stacked coupling transformer is shown in Fig. 9, whereLe1 and Lb2 consist of primary and secondary inductors,respectively. The transformer inverting configuration is imple-mented to form a feed-forward path boosting the transcon-ductance gm2 input stage. All EM effects from eddy currentsubstrate loss to frequency-dependent metal loss are consid-ered in the design process of the transformer.

In order to reduce the parasitic loss effects at highfrequency, the stacked transformer is realized with the topmetal layers M6 and M5, which are the thickest and thefarthest from the substrate, thus reducing losses. The qualityfactor and self-resonance frequency for both Le1 and Lb2remain almost identical. A high quality factor (Q) forthe transformer inductances is needed to reduce its noisecontribution into the balun-LNA structure.

GEHA et al.: WIDEBAND LOW-POWER-CONSUMPTION 22–32.5-GHz 0.18-μm BICMOS ACTIVE BALUN-LNA 543

Fig. 9. Stacked transformer layout structure and its schematic. Port (1, −1):M6. Port (2, 3): M5.

For the optimal magnetic coupling between transformerconductors, the metal width for the microstrip transmissionlines forming the transformer is set to the smallest possible(7.5μm) constrained by the ohmic losses, the dc current, andthe quality factor. The narrower the conductor dimensionswidth the higher the magnetic coupling between the trans-formers turns. However, increasing the metal width leads tohigher parasitic capacitance losses to the substrate.

The coupling coefficient, K , for the stacked transformeris limited by the process technology due to metal thicknessand minimum layers spacing as well as the optimal turn’sratio at mm-wave frequency. Section II-C states clearly thebenefits and limitations of increasing the turn ratios for thestacked transformer. Thus, the stacked transformer is designedwith 1:1 turn’s ratio. Le1 and Lb2 inductances are 82 and120 pH, respectively. A coupling coefficient, K , equal to 0.34is achieved in the band of interest. Fig. 10 shows the EMsimulations results of the transformer inductances and thecoupling coefficient. These parameters remain almost constantin the frequency range of interest. This is because the self-resonance frequency of the transformer is at higher frequency.

IV. SIMULATION AND EXPERIMENTAL RESULTS

The wideband balun-LNA was fabricated using 0.18-μmBiCMOS technology from Tower Jazz Semiconductor [21].Fig. 11 shows the die micrograph of the balun-LNA, where thetotal area is 0.46 mm2 excluding the RF and dc pads. On-wafermeasurements were done using RF differential probes (G-S-G-S-G) for input and outputs. The use of RF differential inputprobe is necessary for calibration purposes using CascadeMicrotech, Inc Impedance Standard Substrate. Although anRF differential probe is used at the input, the input signalis fed into only one port. Also, a 6-pin dc probe is used toprovide the dc biasing. The balun-LNA core consumes 5 mAfrom 1.8 V supply.

Fig. 12 shows the measured and simulated input returnlosses under single (S11) and differential (Sss11) output ter-minations for the balun-LNA, respectively. Measured S11 andSss11 are larger than 8.7 and 10.2 dB, respectively, for theentire operating frequency range of 22–35 GHz and up to40 GHz. Fig. 13 shows the measured and simulated output

Fig. 10. Inductance values, Le1 andLb2, and coupling coefficient, K , forstacked transformer using IE3D.

Fig. 11. Die photograph of the balun-LNA with chip area, including dc andRF pads of 0.8 mm × 0.9 mm.

Fig. 12. Measured and simulated S11 of the proposed balun-LNA. Sss11 isthe measured input return loss for differential load.

return losses S22, S33, and Sdd22. Measured S22 is better than9 dB from 22–29 GHz and S33 is larger than 7.5 dB from23.5–27.4 GHz. Furthermore, the measured differential return

544 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 65, NO. 2, FEBRUARY 2017

Fig. 13. Measured and simulated S22 and S33 of the proposed balun-LNA.Sdd22 is the measured differential output return loss.

Fig. 14. Measured and simulated S21 and S31 for the balun-LNA. Sds21 isthe measured differential output power gain.

Fig. 15. Stability factor of the proposed balun-LNA.

loss Sdd22 is better than 9 dB from 23 to 27.7 GHz. Theshifting of the return loss responses at the outputs of thebalun-LNA is mainly due to the process variations of thesmall metal–insulator–metal output capacitances, as well asthe inaccurate models of those and the parasitic inductancescoupling to the substrate. Consequently, the measured powergains for the balun-LNA (S21, S31, and Sds21) shift to 26.8, 27,and 26.8 GHz, respectively, as seen in Fig. 14, which showsS21, S31, and Sds21 achieving a gain of 15.6, 15.4, and 18.6 dB,respectively. This represents a measured differential gain boost

Fig. 16. Measured and simulated noise figures of the proposed balun-LNA.

of 2.0 and 2.4 dB for S31 and S21 compared with simulations.As expected, the measured differential power gain, Sds21, is3 dB higher than the measured single-channel S21 and S31. Therelatively fast gain drop on S21 around 30 GHz is particularlydue to the higher inductors/interconnects’ parasitic resistancelosses in the signal path of the S21 channel, which werenot accounted for accurately in the models. The measured3-dB bandwidths for S21 and Sds21, and S31 are 7.6 and11.5 GHz, respectively. A 3.9-GHz bandwidth differencebetween S21 and S31 is mainly due to asymmetric signal pathsfrom input to outputs and unequal parasitic capacitances tothe substrate. Fig. 15 shows the measured stability of theproposed balun-LNA in term of the stability parameter μ [22],which is derived from the measured S-parameters. The balun-LNA is unconditionally stable for both channels across the22–32.5-GHz bandwidth according to μ(s) > 1. The measuredNFs for both channels are shown in Fig. 16, where the NFsbetween input port 1 and output port 3 (NF31) and input port 1and output port 2 (NF21) vary from 4.5 to 5.5 dB and from 4.6to 6 dB, respectively. The discrepancies between the channels’measured and simulated NFs are associated with the channels’2.4-dB gain boost and shift in the first half of the frequencyspectrum (22–27 GHz). In the second half (27–35 GHz),NF21 experiences higher NF particularly due to the fastchannel gain drop. In the case of a differential to singleended balun applied at the output of the proposed balun-LNA,a much lower NF can be achieved due to common modenoise cancellation. The measured single-ended-to-differentialoutput gain and phase imbalances are defined as GEds =Mag(S21/S31) and PEds = Phase(S21/S31), where 1 and 2,and 3 are the single-ended input and differential output ports,respectively, and shown in Fig. 17. In addition, the measuredsingle-ended-to-single-ended gain and phase imbalances aredefined as GEss = Mag(S21/S31) and PEss = Phase(S21/S31),where 1 and 2, and 3 are the input and output single-endedports, respectively, and are also reported in Fig. 17. Fordifferential and single output port termination, the gain andphase mismatches from 20 to 32.5 GHz are 2.4 and 1.8 dB,and 9° and 12°, respectively. However, the differential andsingle port termination gain mismatch can reach 4.02 and5.5 dB at 35 GHz. The smallest gain and phase errors occuraround 27 GHz, indicating a near-perfect balance around that

GEHA et al.: WIDEBAND LOW-POWER-CONSUMPTION 22–32.5-GHz 0.18-μm BICMOS ACTIVE BALUN-LNA 545

TABLE II

PROPOSED BALUN-LNA COMPARISON WITH EXISTING LNA DESIGNS

frequency. Away from 27 GHz, the unbalance between thetwo channels is degraded electrically and physically as canbe expected from the schematic and photograph in Figs. 1and 11, respectively, causing the fast phase variation corre-spondingly. The measurements and simulations in a single-to-differential mode configuration of the input referred 1-dBpower compression points (Pin1dBsd), input referred third-order intercept points (IIP3sd), and input referred second-orderintercept points (IIP2sd) for the balun-LNA are shown in Fig.18. Measured Pin1dBsd, IIP3sd, and IIP2sd higher than −14.6,−5.7, and 42 dBm across 22–35 GHz are achieved in theoutput differential mode setup, respectively. The discrepanciesbetween the simulated and measured IIP2sd and IIP3sd for thebalun-LNA are mainly attributed to the unbalanced differentialgain and phase errors and simulated models. The performanceof the proposed wideband balun-LNA is shown in Table IIin comparison with other LNA designs operating at mm-wavefrequencies. These results confirm that the balun-LNA exhibitsgood differential property, high power gain, low NF, verycompetitive linearity, and the lowest power consumption inthe K/Ka-band of operation. The figure of merit (FOM) in[15] is evaluated as

FOM = IIP3,av[mW]· Gain[abs] ·BW3 dB[GHz] · fcenter[GHz](NFav −1) [abs] ·PDC[mW] · f 2

T [GHz]2

(22)

where IIP3,av is the average input-referred third-order inter-cept point, Gain is the absolute power gain, BW3 dB is the3-dB bandwidth, fcenter is the center frequency, NFav isthe average noise figure, PDC is the dc power, and fT is thecorner frequency technology. This FOM is used to comparethe proposed balun-LNA design with others, as reported inTable II. The balun-LNA’s FOM shows a design performancefactor almost 1.5 better than the previously reported results inCMOS and BiCMOS technologies.

Fig. 17. Measured GEsd, GEss, PEsd, and PEss.

Fig. 18. Measured and simulated differential Pin1dB, IIP3, and IIP2 for theproposed balun-LNA.

V. CONCLUSION

A wideband 22–32.5-GHz balun-LNA is implementedusing 0.18-μm SiGe BiCMOS technology. The balun-LNAstructure implemented new linearity technique based onGm-constant approach and utilized coupling staked

546 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 65, NO. 2, FEBRUARY 2017

transformer to improve power and noise efficiency. Analyticalexpressions for the wideband input matching impedance,linearity, NF, and stability were developed to highlightthe design tradeoffs. The gain and phase mismatches forthe frequency range of 20–32.5 GHz are 1.8 dB and12° respectively. Power gains of 15.6 and 15.4 dB, 3-dBbandwidths of 7.6 and 11.5 GHz, NFs of 4.5–5.5 and4.6–6.0 dB, and linearity better than −5.7 dBm are achievedbetween the input and two outputs. The balun-LNA consumesonly 5-mA dc current from 1.8 V supply and having an activearea of 0.46 mm2.

APPENDIX

The noise analysis presented in this Appendix is based onthe noise model, as shown in Fig. 6. Before determining theinput referred voltage noise due to the base and collector shotcurrents of transistors Q1 and Q2, we had to solve various

circuits’ impedances affected by the transformer behavior.From the noise model, Zx is the impedance looking from thebase of transistor Q2 into the transformer. Similarly, Zc is theimpedance at the collector of Q1. Z M is the impedance due toMiller effect at transistor Q1. Furthermore, Ze and Zs are theimpedances looking at the emitter and from the base into thesource generator of Q1, respectively. Applying KCL analysisyields (A1)–(A5) as follows:

Zx ≈ Z2||[

Rbx2 + sLb2

+ (K n) sLe1 (sLb2 + Z1 + Z2) (sCbe + gm1)

gm1 Z1 − (K n) sLe1 (sCbe + gm1)+ Z1

]

(A1)

v2ni,Q1

i2n,c1

≈∣∣∣∣∣∣⎡⎣1 − Ze

Ze +(

resCbere+1

)(1/β)

(ZM Zs

ZM +Zs

)⎤⎦ 1

gm1

∣∣∣∣∣∣2

(A8)

v2ni,Q1

i2n,b1

≈∣∣∣∣∣∣(

ZM ZsZM +Zs

)||((

βresCbeβre+1

)+ β Ze

)+ Ze

Ze+(

resCbere+1

)+(1/β)

((Z M Zs

Z M +Zs

)) 1gm1

∣∣∣∣∣∣2

(A9)

v2ni,Q2

R2bx2

≈∣∣∣∣∣∣[

Z2

Z2 + Zx

]⎡⎣ 1(

gm1gm3

+ nK)⎤⎦∣∣∣∣∣∣2

(A10)

v2ni,Q2

i2n,b2

≈∣∣∣∣∣∣⎡⎣(

Z2 Zx

Z2 + Zx

)+

(RLe2 + sLe2

)(RLe2 + sLe2

)+(

re2sCbe2re2+1

)+ (1/β)

(Zx

sC p2 Zx +1

)⎤⎦[

1gm1gm3

+ nK

]∣∣∣∣∣∣2

(A11)

v2ni,Q2

i2n,c2

≈

∣∣∣∣∣∣∣∣∣∣

⎡⎣1 −

⎡⎣ RLe2 +s Le2(

RLe2 +s Le2+(

re2sCbe2re2+1

))+(1/β)

((Rbx2+Zx )

sCp2(Rbx2+Zx )+1

)⎤⎦⎤⎦

·[

1gm1gm3

+nK

]

∣∣∣∣∣∣∣∣∣∣

2

(A12)

�1 (ω) ≈ 1

2Rs

∣∣∣∣∣∣⎡⎣1 − Ze

Ze +(

resCbere+1

)(1/β)

(ZM Zs

ZM +Zs

)⎤⎦∣∣∣∣∣∣2

(A13)

�2 (ω) ≈ 1

2β Rs

∣∣∣∣∣∣(

ZM ZsZM +Zs

)||((

βresCbeβre+1

)+ β Ze

)+ Ze

Ze+(

resCbe re+1

)+(1/β)

((Z M Zs

Z M +Zs

))

∣∣∣∣∣∣2

(A14)

�3 (ω) ≈Rbx2

∣∣∣[ Z2Z2+Zx

]∣∣∣2Rs

(A15)

�4 (ω) ≈ 1

2β2 Rs

∣∣∣∣∣∣⎡⎣( Z2 Zx

Z2 + Zx

)+

(RLe2 + sLe2

)(RLe2 + sLe2

)+(

re2sCbe2re2+1

)+ (1/β)

(Zx

sC p2 Zx +1

)⎤⎦∣∣∣∣∣∣2

(A16)

�5 (ω) ≈ 1

2β2 Rs

∣∣∣∣∣∣⎡⎣1 −

⎡⎣ RLe2 + sLe2(

RLe2 + sLe2 +(

re2sCbe2re2+1

))+ (1/β)

((Rbx2+Zx )

sC p2(Rbx2+Zx )+1

)⎤⎦⎤⎦∣∣∣∣∣∣2

(A17)

GEHA et al.: WIDEBAND LOW-POWER-CONSUMPTION 22–32.5-GHz 0.18-μm BICMOS ACTIVE BALUN-LNA 547

where Z1 and Z2 are defined in Section II-A

Zc ≈ Z1||[

sLb2

+ (K n) sLe1 (sLb2 + Z1 + Z2) (sCbe + gm1)

gm1 Z1 − (K n) sLe1 (sCbe + gm1)+ Z2

]

(A2)

Zμ ≈ [Zc] + (1/sCbc)

Z M ≈(

Zμ

(1 − Av)

)(A3)

where Av is the voltage gain of the balun-LNA

Ze ≈ sLe1

[1 + (K n) (gm1 Z1 − (K n) sLe1 (sCbe + gm1))

(sLb2 + Z1 + Z2) (sCbe + gm1)

]

(A4)

Zs ≈(

Rbx + RLb + sLb +(

Rs

sCpad Rs + 1

)). (A5)

The base and collector current shot noises for transistorsQ1 and Q2 are given by

i2n,b1,2 = 2q IB1,2 (A6)

i2n,c1,2 = 2q IC1,2 (A7)

where qis the electron charge constant, and IB1,2 and IC1,2are the collector and base currents for transistors Q1 and Q2,respectively.

The input referred voltage noise due to the base and collec-tor currents shot noises of transistors Q1 and Q2, includingthe parasitic base resistance Rbx2, is shown in (A8)–(A12),whereβ is the current gain of Q1 and Q2

Now, taking the outcomes from (A8) to (A12) and nor-malize it to the source generator impedance Rs results in(A13)–(A17). as shown at the bottom of the previous page,�1 (ω) − �5 (ω)is the total equivalent input referred voltagenoise power, as shown in (14)

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Chadi Geha (S’10) received the B.Sc., M.S.E.E.,and Ph.D. degrees in electrical engineering fromTexas A&M University, College Station, TX, USA,in 2005, 2008, and 2015, respectively.

From 2007 to 2008, he was a Test Intern Engineerwith Broadcom Inc., San Jose, CA, USA. He iscurrently a Lecturer/Researcher with Texas A&MUniversity. His current research interests includeRFIC transceiver systems architecture, phased arraysystems, and IC circuits design for microwave andmillimeter-wave frequencies.

Cam Nguyen (F’05), photograph and biography not available at time ofpublication.

Jose Silva-Martinez (SM’98–F’10) was born inPuebla, México. He received the M.Sc. degree fromthe Instituto Nacional de Astrofísica Optica y Elec-trónica (INAOE), Puebla, in 1981, and the Ph.D.degree from the Katholieke Univesiteit Leuven,Leuven, Belgium, in 1992.

In 1993, he joined the Electronics Department,INAOE, where he was the Head of the ElectronicsDepartment from 1995 to 1998. He was also aCo-Founder of the Ph.D. Program on Electronics in1993. He is currently a Texas Instruments Professor

and an Associate Department Head for Graduate Studies Affairs with theDepartment of Electrical and Computer Engineering, Texas A&M University,College Station, TX, USA.